Direct-current-coupled transistor power amplifiers

ABSTRACT

A transistor power amplifier is provided with direct-current-coupling between stages and to the loudspeaker. The output stage is push-pull single-ended and includes two transistors connected in series at a midpoint of the stage. A split power supply has a center-tap connected to one output terminal and the other output terminal is connected to said output stage midpoint. Amplification and drive stages are direct-current-coupled in cascade to the output stage. A direct-current negative feedback network extends from the output stage to an amplification stage to maintain the output terminals at substantially the same quiescent potential to prevent direct-current flow through the voice-coil of the loudspeaker.

CROSS-REFERENCES

This application is related to my prior applications listed below:

"Transistor Power Amplifiers," Ser. No. 270,011, now U.S. Patent No.3,281,535, filed Apr. 2, 1963; "Transistor Power Amplifiers and FeedbackSpeaker Systems Embodying Same," Ser. No. 311,732, .Iadd.now U.S. Pat.No. 3,372,342, .Iaddend.filed Sept. 26, 1963; "Transistor PowerAmplifiers and Feedback Systems Embodying Same," Ser. No. 388,399, filedAug. 10, 1964; "Transistor Power Amplifiers for High Fidelity MusicReproduction," Ser. No. 563,586, filed July 7, 1966.[...]..Iadd.;"Transistor Audio Amplifier with Power-Division Output Stages", Ser. No.501,515, filed Oct. 22, 1965, now abandoned. "Transistor PowerAmplifiers With Protective Circuit Means", Ser. No. 473,981, filed July22, 1965. .Iaddend.

The present application is a continuation-in-part of said applications.Iadd.Ser. No. 501,515, .Iaddend.Ser. No. 311,732; Ser. No. 388,399 nowabandoned; .Iadd.Ser. No. 473,981 now abandoned .Iaddend.and Ser. No.563,586. Said application Ser. No. 388,399 is in turn acontinuation-in-part of said application Ser. No. 270,011 now U.S. Pat.No. 3,281,535.

FIELD OF THE INVENTION

This invention relates to transistor power amplifiers for high-fidelitymusic reproduction and other applications where high power output andinaudible amplifier distortion under rigorously critical listeningconditions are required.

DESCRIPTION OF THE PRIOR ART

In the prior state of the art, the advent of transistors, with theirability to function efficiently when loaded by the relatively lowimpedance of a loudspeaker, made feasible the elimination of the outputtransformer from audio amplifiers. Since the output transformer was byfar the most expensive component and generated the most distortion,phase shift and feedback instability, its elimination would haveprovided a substantial advance if it were not for the fact that priortransistor amplifier circuits substituted other components which werealmost as disadvantageous.

More specifically, one widely adopted design approach provides directcoupling between the output stage and the loudspeaker, but utilizes adriver transformer to drive the output transistors. The primary of thedriver transformer is capacitor-coupled to ground. Another prior artsolution involves direct coupling the drive stage to the output stagebut coupling the latter to the loudspeaker through a large electrolyticcapacitor. Both of these circuit types frequently employ interstagecoupling capacitors as well.

Instead of utilizing these reactive coupling components, the amplifiercircuit of the present invention provides direct coupling between theseveral stages, to the loudspeaker, and throughout the feedback loop. Inaddition to the obvious advantages of reduced cost and greaterreliability due to the elimination of transformers and couplingcapacitors, the primary advantage of the present direct-coupled circuitresides in the elimination or substantial reduction of the followingforms of distortion which are generated by or inadequately reduced bythe conventional circuits:

(1) Nonlinear distortion.--Driver transformers and coupling capacitorsproduce phase shift at both frequency extremes so as to prevent theapplication of a large amount of feedback with stability. Nonlineardistortion, including both harmonic and intermodulation distortion dueto nonlinear transistor transfer characteristics, is reduced in anamount proportional to the amount of feedback. The restriction to amoderate amount of feedback, necessary for stability in the conventionalcircuits, results in a substantial residuum of nonlinear distortion.

(2) Oscillatory response.--At low frequencies this is known as"breathing" and at high frequencies as "ringing." In either case, thecomplex poles of the transfer function provide a spurious oscillatoryresponse if the amount of feedback is excessive in view of the phaseshifts provided by the reactive coupling components of the conventionalcircuits. The designer of the latter is faced with the dilemma ofrisking this form of distortion or settling for a substantial amount ofnonlinear distortion by utilizing only a moderate amount of feedback toobtain a greater stability margin.

(3) Overload recovery.--At overload peaks when one or more transistorsgo into saturation, it is usual for at least one interstage couplingcapacitor to pass a large unidirectional surge of current so as tochange its state of charge. After the overload signal terminates ittakes the capacitor a finite time to recover its original chargecondition. During this recovery period the bias on the succeedingamplification stage is usually sufficiently disturbed by the potentialapplied thereto by the capacitor so as to cause either radicaldistortion or even complete cutoff known as "blocking."

(4) Crossover distortion.--Although inherent in any transistor push-pulloutput stage operating near the Class B bias mode, this type ofdistortion is reduced approximately in proportion to the amount offeedback, provided the distortion is not so severe as to reduce theamplifier gain to the point where the feedback is ineffective. As in thecase of nonlinear distortion, the phase shifts generated by the reactivecoupling components of the conventional circuits prevent the applicationwith stability of a large amount of feedback and hence the residualcrossover distortion is usually excessive.

(5) Transient distortion.--Although this term is sometimes applied tovarious forms of spurious response to nonsinusoidal period signals suchas square waves, it is intended here to refer to distortion arising fromnon-periodic pulse signals. One form produces premature clipping at therise of a tone burst due to a disturbance in the bias condition of anearly stage and produced by a low-frequency instability triggered by theburst. Another form occurs during the recovery from a heavy pulse havinga substantial direct-current component which changes the chargecondition of a coupling capacitor, usually the output capacitor whichcouples to the speaker rather than an interstage capacitor as in thecase of overload recovery distortion.

(6) Bias drift.--Varying thermal and load conditions and changes incomponent parameters may cause a drift of the quiescent operating pointof one or more stages, resulting in premature clipping at power levelslower than rated. The use of reactive coupling in the conventionalcircuits usually prevents the application of a large amount of overalldirect-current feedback which would counteract any tendency of the biaspoint to drift from the optimum value.

(7) Transformer crossover transients.--If the two halves of a drivertransformer primary winding are not perfectly coupled, crossoverswitching transients in the form of notches or ringing will occur,particularly when abetted by carrier storage at high frequencies.Trifilar windings increase the coupling and thus reduce thesetransients, but do not completely eliminate them.

(8) Reactive coupling loads.--The use of an output coupling capacitorresults in an elliptical load-line for the amplifier at low frequencies,and the primary inductance of a driver transformer provides a similarreactive load for the drive stage. In response to low-frequencytransient signals the operating point of the output stage may enterregions of simultaneously high voltage and current so as to result in"second breakdown" causing the destruction of the output transistors.

SUMMARY OF THE INVENTION

It is therefore a primary object of the present invention to providenovel transistor power amplifier circuits wherein the pre-drive, driveand output stages are D.C.-coupled to each other and the output stage isD.C.-coupled to the loudspeaker, so as to eliminate the seriousdisadvantages inherent in the use of driver transformers and outputcoupling capacitors. Prior art circuits having the last two or threestages D.C.-coupled in this manner have heretofore been unsuccessfulbecause they are deficient in one or more of the following respects:

(1) There was an inability to maintain D.C. balance at the amplifieroutput terminals under varying ambient temperatures, line voltages andoutput power loads. Any substantial D.C. unbalance produces a D.C.current in the loudspeaker voice-coil thereby biasing the cone offcenter and increasing the speaker distortion.

(2) The power supply ripple was injected into an early stage of theamplifier thereby producing an excessive amount of hum.

(3) The power and distortion characteristics did not compare favorablywith other circuits.

(4) Some of the circuits were complex and critical and therefore costlyand unreliable.

(5) Failure of an output transistor was likely to damage the loudspeakerso as to require factory replacement of the cone and voice-coil atsubstantial expense.

(6) Although measuring extremely well on distortion meters in laboratorybench tests they generated audible distortion when subjected to musicsignals in critical listening tests.

It is therefore a further object of the present invention to obviate theabove-noted defects heretofore prevalent in D.C.-coupled amplifiercircuits.

The attainment of practical, and properly functioning circuitry whereinthe pre-drive, drive and output stages are D.C.-coupled to each otherand to the loudspeaker provides an amplifier having phase shiftcharacteristics which permit the use of new and more effective feedbacktechniques and impedance matching arrangements.

In one embodiment an inner negative feedback loop raises thehigh-frequency cutoff of the amplifier to a predetermined frequency, andpreceding the inner loop is a preamplifier stage having a high-frequencycutoff substantially lower than said predetermined frequency. An outernegative feedback loop includes both the amplifier and the preamplifierstage. The result is an unusually large amount of feedback withoutinstability.

Other objects and advantages are either inherent in the circuitrydisclosed herein or will become apparent to those skilled in the art asthe detailed description proceeds in connection with the accompanyingdrawings wherein:

DESCRIPTION OF THE DRAWINGS

FIG. 1 is an amplifier circuit embodying the invention;

FIG. 2 shows a modified form of the invention which uses only a singlezener diode and a non-split auxiliary power supply, and which alsoincludes an additional four-transistor drive stage having twotransistors driven by the output stage through feedback voltage dividersenergized by the latter;

FIG. 3 is a schematic circuit diagram of a double-loop feedback systemincorporating an amplifier in accordance with the present invention;

FIG. 4 shows another double-loop feedback system wherein the secondpreamplifier stage is D.C.-coupled to the amplifier;

FIG. 5 shows two amplifiers each in accordance with the presentinvention and connected in a stereo mode so as to constitute twoindependent channels;

FIG. 6 shows the same two amplifiers connected in a parallel mode so asto match the relatively low impedance of a single four-ohm loudspeaker;

FIG. 7 shows the same two amplifiers connected in a series mode to forma full-bridge circuit for driving a single loudspeaker of high impedancesuch as sixteen ohms.

FIG. 8 shows a modified form of the invention wherein a zener diodeprovides a source of reference potential for the base of the firstcommon-emitter stage;

FIG. 9 shows another modified form of the invention wherein the groundis a source of reference potential and an emitter-follower stagemaintains the base of the first common-emitter stage at a fixedquiescent potential;

FIG. 1a shows still another modified form of the invention wherein thesource of reference potential is at ground level;

FIG. 2a shows a modification of the first two stages of the circuit ofFIG. 1a;

FIG. 3a shows another modification of said first two stages of thecircuit of FIG. 1a;

FIG. 4a shows another modified form of the invention wherein eachcommon-emitter stage comprises a symmetrical pair of emitter-coupledtransistors; .[.and.]..Iadd.

FIG. 4b shows a modified form of the invention and consisting of asubcombination of the circuit of FIG. 4a; and .Iaddend.

FIG. 5a shows a widely-used prior art circuit utilizing an outputcoupling capacitor.

DETAILED DESCRIPTION

Referring now to the drawings in more detail and first to FIG. 1thereof, there is shown an amplifier circuit in accordance with thepresent invention and comprising a voltage amplification or pre-drivestage including transistor Q1, a complementary symmetry push-pull drivestage including transistors Q2 and Q3, and a push-pull power outputstage including transistors Q4 and Q5. Each stage is D.C.-coupled to thenext stage and the output stage is in turn D.C.-coupled to theloudspeaker S. The pre-drive transistor Q1 is operated common-emitter inClass A. The drive and output stages operate as emitter-followerDarlington pairs and are preferably biased in Class AB with sufficientquiescent current to minimize crossover distortion.

Transistor Q1 is provided with a base biasing circuit comprising theseries arrangement of diode D1, resistor R1, resistor R2, and a variableresistor in the form of a potentiometer P. The base of transistor Q1 isconnected to the junction of resistors R1, R2. The emitter of transistorQ1 is provided with a bias resistor R3 which is bypassed by a capacitorC2. The collector of transistor Q1 is connected to one end of a diode D2having its opposite end connected to collector load resistors R5, R6'.Capacitor C1 has one end connected to output terminal O1 through leads15, 16 and its opposite end connected to the junction of resistors R5,R6' to provide bootstrapping in the conventional manner.

The battery symbols B2 and B'2 designate the half-sections of aconventional auxiliary power supply having its center-tap grounded asshown. A resistor R6 has one end connected to the positive terminal ofthe positive half-section B2 of the power supply and its opposite end isconnected to the positive end of a zener reference diode Z1. Thenegative end of the latter is connected to ground. The lower ends ofdiode D1, resistor R3 and bypass capacitor C2 are connected to thepositive end of zener diode .[.D1.]. .Iadd.Z1.Iaddend..

A resistor R7 has one end connected to the negative end of the negativehalf-section B'2 of the auxiliary power supply and its opposite end isconnected to the negative end of a second zener reference diode Z2having its positive end connected to ground. The upper end of collectorload resistor R5 is connected to the negative end of zener diode.[.D2.]. .Iadd.Z2.Iaddend..

The potential of the positive terminal of power supply positive sectionB2 is more positive than the breakdown voltage of zener diode Z1 so thatthe latter is maintained in its breakdown state with its positive end ata substantially fixed predetermined voltage with respect to ground.Similarly, the negative terminal of power supply negative section B'2 ismaintained at greater negative potential than the breakdown voltage ofzener diode Z2 so that the latter is maintained in the breakdown statewith its negative end at a fixed predetermined negative voltage withrespect to ground. Therefore the base, emitter and collector circuits oftransistor Q1 are connected to potential sources at fixed predeterminedvoltages which remain substantially invariant under different conditionsof ambient temperature, line voltage and amplifier load.

The collector of transistor Q1 is D.C.-coupled by lead 11 to the base ofthe lower NPN transistor Q2 of the complementary drive stage and theupper end of diode D2 is similarly D.C.-coupled by lead 12 to the baseof the upper PNP transistor Q3 of said drive stage. Bias resistors R9,R10 extend from the respective emitters of transistors Q2, Q3 to acommon junction connected by lead 15 to the output terminal Q1. Thecollector of the lower NPN transistor Q2 is D.C.-coupled by lead 13 tothe base of the lower output transistor Q4 and the emitter of the upperPNP drive transistor Q3 is D.C.-coupled by lead 14 to the base of theupper output transistor Q5.

Resistor R8 has one end connected to the collector of drive transistorQ2 and its opposite end is connected to the positive terminal of thepositive half-section B1 of a main split power supply having a negativehalf-section B'1 with a negative terminal connected to the collectors ofboth upper drive transistor Q3 and upper output transistor Q5. Thecenter-tap of the main power supply B1, B'1 is grounded as shown andconstituted the grounded output terminal O2 whereby the speaker S may beconnected with its voice-coiled terminals D.C.-coupled to the respectiveamplifier terminals O1, O2.

The emitter of the lower output transistor Q4 is provided with a biasresistor R11 extending to the positive terminal of the positive powersupply section B1 and its collector is connected to output terminal O1.The emitter of upper output transistor Q5 is similarly provided with abias resistor R12 extending to output terminal O1.

The two signal input terminals are indicated at I1, I2. Input terminalI1 is connected directly to the base of transistor Q1 whereas inputterminal I2 is grounded as shown. Transistor Q1 operates as a Class Acommon-emitter amplifier stage and the potential of its collector variesin response to a signal fed to input terminals. I1, I2. Diode D2provides a temperature-compensated bias for drive transistors Q2, Q3 inthe conventional manner.

Due to the small voltage drop across diode D2 the potential of lead 12will be maintained a fraction of a volt more negative than the collectorof transistor Q1 as the collector swings in response to a signal input.The complementary drive stage Q2, Q3 provides both phase inversion andemitter-follower operation. That is, since the emitter of a transistoroperating in the active region remains within a fraction of a volt ofthe base, it will be seen that the potential of the junction of emitterresistors R9, R10, and hence that of the output terminal O1, will bemaintained intermediate the respective potentials at the opposite endsof diode D2 as the latter swings up and down with the collector oftransistor Q1. Thus the potential of amplifier output terminal O1 willbe maintained within a fraction of a volt of the potential of thecollector of transistor Q1 so as to provide the low distortion ofemitter-follower operation.

In view of the D.C.-coupling of the amplifier to the loudspeaker S, itis imperative that the output terminals O1, O2 be maintained atsubstantially the same D.C. potential, both under quiescent conditionand at various levels of power output, and irrespective of variations inambient temperature and line voltage. Any D.C. unbalance between outputterminals O1, O2 will cause a direct current to flow through thevoice-coil of loudspeaker S. This in turn will cause the cone ofloudspeaker S to be biased off center. If the direct current and theresultant biasing are substantial, the nonlinear distortion generated bythe loudspeaker will be seriously increased.

Output terminal O1 is maintained at substantially the same potential asoutput terminal O2, that is, at ground potential, by two feedbackarrangements. The first comprises the emitter bias resistor R3 oftransistor Q1 which is made highly effective by zener diode Z1. Thebreakdown voltage of zener diode Z1 is selected so as to besubstantially larger than the quiescent collector-to-emitter voltage oftransistor Q1 so that the extra voltage may be dropped across resistorR3. The resistance of the latter may thus be many times larger than iscustomary or would be permissible in the absence of zener diode Z1.

For example, in conventional prior art circuits of thecomplementary-symmetry type wherein the lower ends of resistors R3, R8and R11 are connected to a common ground, the resistor in the relativeposition of R3 is generally about a few hundred ohms. Any attempt toincrease this resistance substantially will cause a correspondingreduction in the available voltage swing of the collector of Q1 andhence in the maximum power output capability of the amplifier. However,if the breakdown voltage of zener diode Z1 is selected so as to be apredetermined amount (e.g., about ten volts) greater than the voltage ofthe positive terminal of the main power supply section B1 then resistorR3 may have this voltage difference dropped across it. Hence, for thesame emitter current, resistor R3 may be of a magnitude of severalthousand ohms, that is, approximately ten times larger than in theconventional prior art circuits, without any reduction in the collectorvoltage swing or the amplifier maximum power output.

Since resistor R3 provides D.C. feedback which stabilizes the collectorcurrent of transistor Q1 and hence the potential of the collector oftransistor Q1, by increasing the magnitude of this resistor ten-fold,the feedback is increased by the same amount and hence the stability ofthe D.C. potential of output terminal O1 is increased by a like amount.

The second feedback arrangement for the latter purpose is provided bythe base bias circuit including elements .[.D1.]. .Iadd.Z1.Iaddend., R2and P. The potentiometer P and resistor R2 together with leads 15 and 16provide a D.C. feedback path extending from output terminal O1 to thebase of transistor Q1. This feedback is degenerative so that itcounteracts any tendency of output terminal O1 to vary in D.C.potential. Potentiometer P further provides a convenient arrangement forinitially adjusting the potential of output terminal O1 so that it is atground potential and thus balanced with respect to the grounded outputterminal O2.

The D.C. feedback provided by potentiometer P and resistors R1, R2 isaided by zener diode Z1 in two important respects. First, thepredetermined invariant potential provided by zener diode .[.D1.]..Iadd.Z1 .Iaddend.constitutes a fixed reference voltage which thefeedback arrangement utilizes as a comparison standard to detect andcorrect errors in the D.C. potential of output terminal O1. Second, thepotential source arrangement of zener diode .[.D1.]. .Iadd.Z1.Iaddend.provides a large potential drop across the lower base biasresistor R1 and hence the latter may be many times larger than isconventional. Since the amount of D.C. feedback is determined by theratio of the magnitude of resistor R1 to the sum of the magnitudes ofresistors R1, R2 and potentiometer P, the greatly increased value of R1results in a substantially larger amount of D.C. feedback and hencegreater effectiveness in maintaining the D.C. potential output terminalO1 fixed at ground level.

Further precision in this regard is provided by diode D1 which serves asa temperature compensating element to counteract the variations in thebase-to-emitter voltage of transistor Q1 with variations in ambienttemperature. The temperature compensating operation of diode D2 iswell-known in the art and similarly counteracts the variation inbase-to-emitter voltage of drive transistors Q2, Q3 and outputtransistors Q4, Q5 with variations in ambient temperature.

It will be seen that variations in amplifier load will have practicallyno effect on the D.C. balance of output terminals O1, O2. That is, asthe load increases the regulation of the main power supply B1, B'1 willcause the voltage across its terminals to decrease substantially due tothe internal impedance of the supply. The center-tap at O2 will remainat the same potential since it is grounded and the respective potentialsat the opposite ends of the power supply will shrink toward ground asthe load on the amplifier is increased.

However, the D.C. potential of output terminal O1 will remainsubstantially unaffected for the following reasons. The potential ofoutput terminal O1 is determined by the potential of the collector oftransistor Q1. Zener diodes Z1, Z2 maintain the lower end of emitterresistor R3 and the upper end of collector resistor R5 at substantiallyconstant fixed voltages irrespective of variations in the load on theamplifier. Hence the D.C. potential of the collector of transistor Q1remains substantially constant, and thus the D.C. potential of outputterminal O1 is maintained at ground potential under varying loadconditions.

For the same reasons, variations in line voltages will not disturb theD.C. balance of output terminals O1, O2 provided that the line voltagebe at least the minimal value required to cause the voltages of theauxiliary power supply sections B2, B'2 to be greater than therespective breakdown voltages of zener diodes Z1, Z2 so that the lattermay operate in their proper breakdown regions.

The D.C.-coupling between the three high level stages and between theoutput stage and the loudspeaker provides several important advantages,the foremost probably being the phase shift characteristics which permitdegrees and techniques of feedback not possible with conventionaltransformer and capacitor coupling arrangements. At low frequencies theD.C.-coupling introduces no phase shift whatever. In addition, therelatively large resistance of emitter resistor R3 further improves thelow frequency stability margin in any feedback arrangement by providingsubstantial degeneration and consequent large gain reduction at extremelow frequencies where the bypass capacitor C2 becomes effectively open.

At high frequencies the large feedback stability margin made possible bythe emitter-follower operation and Darlington pair arrangement of thelast two stages is further augmented by the complete elimination ofstray capacitances due to coupling capacitors and transformers, as wellas leakage inductance of the latter. The feedback techniques madeavailable by these greatly improved phase shift characteristics aredescribed hereinbelow.

Another advantage of the D.C.-coupling arrangement resides in theimproved overlaod recovery characteristics due to the absence of anyinterstage coupling capacitors preceding a high level stage. In prioramplifier circuits having such capacitors, overloading of the stagecaused a large current to flow through the capacitor so as to change itsvoltage. This caused the stage to be improperly biased, and most suchamplifiers generated severe distortion or even blocked completely for ashort period until the capacitor recovered its normal voltage and thebias resumed its proper value.

Still another advantage of the D.C.-coupling arrangement is thereduction of the tendency to second breakdown of the output transistors.The mechanism of second breakdown is not completely understood but isbelieved to result from the simultaneous occurrence in the transistor ofhigh instantaneous voltage and high instantaneous current. It isprobable that in most instances this condition occurs because of thereactive nature of the load at low frequencies when an output couplingcapacitor is employed between the output stage and the loudspeaker.

Still another advantage of the present invention resides in the completeisolation of the ripple of the main power supply B1, B'1 from thevoltage amplification stage comprising transistor Q1. The auxiliarypower supply B2, B'2 has negligible ripple, even under heavy amplifierload, because it supplies a very small current which remains small dueto the Class A operation of Q1. Also, any residual ripple is smoothedout by the filtering action of the zener diodes.

A further advantage of the D.C.-coupling resides in the impedancematching of the amplifier to the load. The usual 1,000 microfarad outputcoupling capacitor has at 20 cycles per second twice the impedance of afour-ohm loudspeaker, and hence the maximum power capability of theamplifier is greatly reduced when driving such a loudspeaker. Thisdefect is entirely obviated by the present D.C.-coupling to the speaker.

Furthermore, the present invention makes feasible a modular arrangementwhereby two amplifier sections may be selectably connected in either aparallel mode to match low impedance speakers (e.g., four ohms) or in aseries mode to match high impedance speakers, such as sixteen ohms.

The price to be paid for the D.C.-coupling arrangement of the disclosedembodiment is rather small in view of the substantial advantages notedabove. The auxiliary power supply B2, B'2 supplies only a fewmilliamperes and hence is relatively inexpensive. It may comprise anextra winding or tap on the power transformer of the main power supplyB1, B'1 so as to obviate the need for an extra transformer. It is to beunderstood that other voltage reference means may be substituted for thezener diodes. For example, there may be utilized other types ofbreakdown components such as gas tubes, or ordinary D.C. batteries, orthe auxiliary power supply B2, B'2 may be of the regulated type.However, the zener diode arrangement disclosed is simpler, lessexpensive and more reliable than these other expedients.

Referring now to FIG. 2, there is disclosed another modified form of theinvention wherein the pre-drive stage is directly connected to the mainpower supply and utilizing only a non-split auxiliary power supply and asingle zener diode.

In more detail, the amplifier of FIG. 2 comprises a first pre-drivestage including a transistor Q1x operating in the Class A common-emittermode. The base of transistor Q1x is connected to the input terminal I1x,the other input terminal I2x being grounded as shown. The base is biasedby resistors R1x, R2x and potentiometer Px. The emitter of transistorQ1x is provided with two bias resistors R3x, R4x. The lower resistor R3xis bypassed by a capacitor C1x whereas the upper resistor R4x isunbypassed, for a purpose to be described. The collector of transistorQ1x is connected to the lower end of a load resistor R5x.

A zener diode Z has its positive end connected to ground and itsnegative end is connected to the upper end of collector load resistorR6x and the upper end of potentiometer Px whereby the collector and thebase bias circuits are energized by a potential source which ismaintained at a predetermined fixed voltage independent of thevariations in amplifier load, line voltage and ambient temperature. Thenegative end of zener diode Z is connected by resistor R7x to thenegative terminal of an auxiliary power supply B2x having its positiveterminal connected to the ground bus G.

The second pre-drive stage comprises an NPN transistor Q2x alsooperating in the Class A common-emitter mode. The base of transistor Q2xis direct-coupled by lead 201 to the collector of the first pre-drivetransistor Q1x. The emitter of transistor Q2x is connected through abias resistor R10x to the negative terminal of the negative section B'1xof the split main power supply indicated schematically by the batterysymbols. The collector of transistor Q2x is connected to the upper endof a temperature compensating bias diode D having its lower endconnected to the series-connected load resistors R8x, R9x. Aconventional bootstrapping capacitor C3x has its lower end connected tothe junction of collector load resistors R8x, R9x and its upper end isconnected through lead 215 to the output bus O.

The circuit of FIG. 2 comprises two drive stages, the first drive stagebeing the complementary symmetry type and including an NPN drivetransistor Q3x and a PNP drive transistor Q4x. Bias resistors R16x, R17xextend from the respective emitters of transistors Q3x, Q4x to outputbus O. The collector of NPN drive transistor Q3x is connected bytransistor R15x to the positive terminal of the positive section B1x ofthe main power supply.

The second drive stage comprises four PNP transistors Q5x, Q6x, Q7x,Q8x. The base of the lowermost drive transistor Q5x is direct-coupled bylead 204 to the collector of transistor Q3x and the base of drivetransistor Q7x is direct-coupled by lead 205 to the emitter oftransistor Q4x. Resistor R18x extends from the emitter of drivetransistor Q5x to the positive terminal of power supply section B1x andresistor R19x extends from the emitter of drive transistor Q7x to theoutput bus O. The collector of transistor Q6x is connected to the outputbus O and the collector of transistor Q8x is connected to the negativeterminal of the negative B'1x of the main power supply.

The potential of the base of drive transistor Q6x is maintained midwaybetween the potentials of .[.ground.]. .Iadd.output .Iaddend.bus O andthe positive terminal of the power supply by a feedback networkenergized by the output of the amplifier and comprising the voltagedivider resistors R11x, R12x arranged in series from the power supplypositive terminal to the output bus O. The base of transistor Q6x isconnected by lead 213 to the junction of voltage divider resistors R11x,R12x. The resistance values of the latter are preferably approximatelyequal.

In a similar manner the potential of the base of drive transistor Q8x ismaintained approximately midway between the potentials of output bus Oand the power supply negative terminal by means of a feedback networkcomprising the voltage divider resistor R13x, R14x extending from outputbus O to the power supply negative terminal. The base of drivetransistor Q8x is connected by lead 214 to the junction of resistorsR13x, R14x. The resistance values of the latter are also preferablyapproximately equal. As a result, the transistors of both the seconddrive stage and the output stage will share approximately equally boththe quiescent D.C. potentials and the varying A.C. potentials as theoutput bus O swings in response to a signal input to the amplifier.

The output stage of the circuit of FIG. 2 comprises four powertransistors Q9x, Q10x, Q11x, Q12x arranged in series. The emitter of thelowermost output transistor Q9x is connected through bias resistor R20xand fuse F1x to the power supply positive terminal and its collector isconnected to the emitter of the next higher output transistor Q10x. Thecollector of the latter is connected to the output terminal O1x to whichthe emitter of output transistor Q11x is also connected through the biasresistor R21x and fuse F2x. The collector of output transistor Q11x isconnected to the emitter of the uppermost output transistor Q12x, thelatter having its collector connected to the power supply negativeterminal.

The second drive stage is direct-coupled to the output stage in thefollowing manner so as to form four Darlington compound pairs. Theemitter of each of the second drive stage transistors Q5x, Q6x, Q7x, Q8xis direct-coupled by one of the leads 207, 208, 209, 210, 212 to thebase of a respective one of the output transistors Q9x, Q10x, Q11x,Q12x. The collector of the first drive stage transistor Q4x is connectedby lead 206 to the collector of the second drive stage transistor Q7xwhich is in turn connected by lead 211 to the collector of outputtransistor Q11x. The collector of drive transistor Q5x is connected bylead 208 to the collector of output transistor Q9x.

Output terminal O1x is directly connected to the midpoint of the outputstage at the collector of transistor Q10x. The other output terminal O2xis grounded as shown. Hence the loudspeaker S is D.C.-coupled to theamplifier so as to provide all of the advantages inherent in thisarrangement and discussed above with respect to the circuit of FIG. 1.In order to maintain the output terminals O1x, O2x at the same D.C.potential so as to prevent the flow of D.C. current in the voice-coil ofthe loudspeaker, there are provided two feedback systems.

The first feedback system arises from the degeneration inherent in theresistors R3x, R4x in the emitter circuit of transistor Q1x. Theseresistors counteract any tendency for the collector current oftransistor Q1x to vary with changes in temperature. The zener diode Zmaintains the upper end of collector load resistor R5x and the upper endof potentiometer Px at a substantially fixed voltage. Therefore thepotential of collector Q1x is maintained substantially constant andhence the D.C. potential applied to the base of the second pre-drivetransistor Q2x remains substantially fixed.

The second feedback system for maintaining output terminals O1x, O2x atthe same D.C. potential comprises a D.C. feedback network extending fromthe output O1x to the emitter of the first transistor Q1x. This networkincludes a feedback resistor R6x, connected in parallel with the usualphase-advance capacitor C2x, and having one end connected to output O1xand its opposite end connected to the emitter of transistor Q1x. Thisfeedback network is degenerative so as to counteract any tendency ofD.C. voltage of output O1x to vary from ground potential. This feedbacknetwork also serves the additional function of providing A.C. negativefeedback for the usual purposes of reducing distortion, improvingfrequency response, and reducing the output impedance of the amplifier.In order to reduce the A.C. feedback to an amount which will provide anadequate stability margin the lower emitter resistor R3x is bypassed bythe capacitor C1x.

Referring now to FIGS. 3 and 4, there are disclosed two embodiments of adouble-loop feedback arrangement made more feasible by the improvedphase shift characteristics due to the D.C.-coupling feature of theamplifier circuits shown in FIGS. 1 and 2. Each of these feedbacksystems comprises an inner negative feedback network extending aroundthe final several stages, and providing, in addition to the usualfeedback advantages of reduced distortion, an improved pole-zeroconfiguration for this portion of the amplifier circuit so as to permitthe application of still more feedback through the outer negativefeedback network. That is, the inner feedback loop provides an addedstability margin for the outer feedback loop.

Referring now to FIG. 3 in more detail, the reference character A2designates a symbol for an amplifier which may be in accordance with thecircuit of FIG. 1. The circuit of FIG. 2 may also be utilized providedthat one of the preamplifier stages be omitted to take into account theextra phase inversion provided by the extra common-emitter pre-drivestage in this amplifier embodiment, as will be explained below.

The double-feedback system of FIG. 3 further comprises a firstpreamplifier stage operating in the Class A common-emitter mode andcomprising a transistor Q17 having its base connected by resistor R45 tothe input terminal 17. Resistor R45 prevents the feedback current fromflowing into the preceding stage (not shown) instead of into the base ofQ17. The other input terminal I8 is connected to ground bus G. Alsoconnected to the latter are a base bias resistor R46 and an emitterresistor R48. The latter is bypassed by a capacitor C19. The other basebias resistor R47 extends to a potential source B₃ +. Extending from thelatter to the collector of transistor Q17 is a load resistor R49.

The collector of transistor Q17 is direct-coupled to the base of atransistor Q18 constituting a second preamplifier stage and having acollector load resistor R51 connected to the potential source B₃ + andan emitter resistor R50 connected to ground bus G. The collector oftransistor Q18 is coupled by capacitor C20 to the input terminal 11e ofamplifier A2.

Said amplifier input terminal 11e may correspond to terminal I1 ofFIG. 1. Amplifier A2 is shown grounded to bus G by lead 123. The outputterminal O1e of amplifier A2 is direct-coupled to speaker terminal T1 ofloudspeaker S by lead 128. Output terminal O1e may correspond to outputterminal O1 of FIG. 1. The other speaker terminal T2 is grounded to busG.

The inner feedback network comprises a feedback resistor R52 in parallelwith a capacitor C21 and connected by lead 124 to the output terminal01e and by lead 125 to the emitter of preamplifier transistor Q18. Theouter feedback network comprises a feedback resistor R53 in parallelwith a capacitor C22 and connected by lead 126 to output terminal 01eand by lead 127 to the base of the first preamplifier transistor Q17.

The transistor and network parameters may be selected so as to provideeither of two general modes of operation. Either the closed-looptransfer function of that portion of the circuit enclosed by the innerfeedback network, hereinafter referred to as the "inner transferfunction," has a substantially higher cutoff frequency than that of apreceding preamplifier stage, or said inner transfer function has asubstantially lower cutoff frequency than all of the precedingpreamplifier stages.

In the first mode of operation, capacitor C21 provides a phase-advanceto improve the pole-zero configuration of the inner transfer function.The inner feedback network raises the high-frequency cutoff of thisportion of the circuit by approximately the amount of the innerfeedback. Transistor Q17 should be of a type having a beta cutofffreqency substantially lower than the cutoff frequency to which theinner transfer function is boosted by the inner feedback network.Alternatively, this roll-off in the first preamplifier stage may beprovided by a filter network such as a resistor and a capacitorconnected in series between the collector of Q17 and ground.

Therefore, as the signal frequency is increased, attenuation will occurinitially only in the first preamplifier stage comprising transistorQ17, and the remaining stages enclosed by the inner feedback networkwill have neither frequency response attenuation nor phase shift to anysignificant degree until the signal frequency is increased to about adecade beyond the beta cutoff frequency of transistor Q17. By this time,transistor Q17 has provided sufficient attenuation so that the outerloop gain is reduced to below unity before the phase shift of the innertransfer function reaches 90°. Since the maximum phase shift in thefirst preamplifier stage is 90°, the overall loop gain will be reducedto below unity before the outer loop phase shift reaches 180°, and hencestability for the outer feedback loop is assured by the increased highfrequency response of the inner transfer function provided by the innerfeedback network.

The alternative mode of operation requires that the upper cutofffrequencies of all preceding preamplifier stages be subtantially greaterthan that of the inner transfer function. In the circuit of FIG. 3 thiscondition is satisfied by providing an early frequency rolloff in atleast one of the stages of amplifier A2 and by selecting for Q17 atransistor having a very high beta cutoff frequency. In this event theinner transfer function provides sufficient attenuation so that theouter loop gain is reduced to below unity before the phase shift of Q17reaches 90°, and hence before the outer loop phase shift reaches 180°.

Still another form of double-loop feedback system is shown in FIG. 4wherein the second preamplifier stage is direct-coupled to the amplifierand the first preamplifier stage is capacitor-coupled to the secondpreamplifier stage, so as to be the reverse of the coupling arrangementof FIG. 3. More specifically, a first preamplifier transistor Q22 isprovided with base bias resistors R70, R80, an emitter resistor R81bypassed by a capacitor C32, and a collector resistor R82 connected to apotential source B₇ -. The base of Q22 is connected to input terminalI13 by resistor R69 and the other input terminal I14 is grounded to busG.

The collector of transistor Q22 is coupled by a capacitor C33 to thebase of a second preamplifier transistor Q23 having an emitter resistorR84 and a base bias resistor R83 connected to ground bus G and acollector load resistor R85 and a base bias resistor in the form ofpotentiometer Ph conected to a potential source B₆ +. The latter ispreferably the junction of the positive end of zener diode Z1 of FIG. 1and resistor R6 in the event that the amplifier A5 of FIG. 4 is in theform of said FIG. 1.

This circuit will be modified so as to eliminate therefrom the base biasnetwork of the pre-drive stage including components D1, R1, R2 and P ofFIG. 1. This bias network is omitted because the collector of transitorQ23 is direct-coupled to the base of the pre-drive stage transistor soas to set the bias of said base. The D.C. potential of output terminalO1h of amplifier A5 if set to ground level by adjustment ofpotentiometer Ph. Amplifier A5 is grounded to bus G by lead 139.Terminal T1 of loudspeaker S is direct-coupled to output terminal O1h bylead 144 and the other speaker terminal T2 is grounded.

The inner feedback network comprises a resistor R86 in parallel with acapacitor C34 and having one end connected by lead 140 to outputterminal O1h and its other end connected by lead 141 to the emitter oftransistor Q23. Because of the D.C.-coupling between transistor Q23 andamplifier A5, the inner feedback network of FIG. 4 serves two functions.In addition to the function of the inner feedback network of FIG. 3 asdescribed above in connection with the latter, the inner feedbacknetwork of FIG. 4 also provides D.C. negative feedback which helps tomaintain the D.C. potential of output terminal O1h at ground level so asto prevent direct current in the loudspeaker voice-coil. The outerfeedback network is similar to that of FIG. 3 and comprises a resistorR87 in parallel with a phase-advance capacitor C35 and connected at oneend to output terminal O1h by lead 142 and the other to the base oftransistor Q22 by lead 143.

Referring now to FIGS. 5 to 7, there is shown the manner in which twoamplifiers in accordance with the present invention may be connectedalternatively in either a stereo mode (FIG. 5) to provide twoindependent channels, or in a parallel mode (FIG. 6) to drive a singlelow-impedance speaker, or in a series mode (FIG. 7) to drive a singlehigh-impedance speaker. This alternative connection capability is madefeasible by the fact that the present invention maintains the D.C.potential of the amplifier output terminals at a constant ground level.

Describing in more detail first the stereo mode connection of FIG. 5,amplifiers A6 and A7 may be identical and may embody any of theamplifier circuits described above. Each amplifier is preceded by apreamplifier stage comprising a transistor Q24 and Q25, respectively.Each of the latter is provided with an emitter resistor R88 and R90 anda collector resistor R89 and R91, respectively. Resistor R91 isconnected to a potential source B₈ - and resistor R89 is connected tothe latter by lead 145. The base of Q24 is coupled by capacitor C36 tothe input terminal I15 and the base of Q25 is similarly coupled bycapacitor C37 to the input terminal I17. The two input terminals I16 andI18 are grounded to bus G. The emitter of Q24 is coupled by capacitorC38 to the input of amplifier A6 and the emitter of Q25 is coupled bycapacitor C39 to the input of amplifier A7. Amplifier A6 is grounded tobus G by lead 148 and amplifier A7 is similarly grounded by lead 149.

The reference letters S and S' indicate respectively the loudspeakersfor the left and right channels. The terminal T1 of loudspeaker S isdirect-coupled by lead 146 to the output of amplifier A6 and theterminal T1a of loudspeaker S' is direct-coupled by lead 147 to theoutput of amplifier A7. The other speaker terminals T2 and T2a aregrounded to bus G. It will thus be seen that in the connectionarrangement of FIG. 5 transistors Q24 and Q25 operate in theemitter-follower mode and that there are two independent channels eachdriving a respective one of the loudspeakers S, S'.

Describing now the parallel connection of FIG. 6, the same components asin FIG. 5 are employed but are connected in a different mode by aswitching arrangement which is not shown since it would be obvious toone skilled in the art view of the disclosed circuit diagram. Inputterminal I19 is coupled both to the base of Q24 by capacitor C36 and tothe base of Q25 by lead 152 and capacitor C37. The other input terminalI20 is grounded.

The terminal T1 of a single loudspeaker S is direct-coupled both to theoutput of amplifier A6 by lead 150 and also to the output of amplifierA7 by lead 151. The other speaker terminal T2 is grounded. It will thusbe seen that the amplifiers A6, A7 are driven in phase from a singlepair of input terminals and are D.C.-coupled in parallel to each otherand to the loudspeaker S.

Describing now the series connection of FIG. 7, the latter differs fromthat of FIG. 6 in only two respects. First, the loudspeaker S has oneterminal T1 direct-coupled by lead 153 to the output of amplifier A6 andits other terminal T2 direct-coupled by lead 154 to the output ofamplifier A7. Second, the input of amplifier is coupled by capacitor C39to the collector of transistor Q25, instead of to the emitter of thelatter as in FIG. 6. The second change provides a phase reversal intransistor Q25 so that the amplifiers A6 and A7 are driven in phaseopposition to form a push-pull full-bridge output circuit for drivingloudspeaker S.

Since amplifiers A6, A7 are connected in parallel in the connection ofFIG. 6, each amplifier will "see" a load impedance of twice theimpedance of loudspeaker S. Hence if the latter is of relatively lowimpedance, such as, for example, four ohms, each amplifier will have aneffective load of eight ohms which is a better match for most powertransistors. However, in the connection of FIG. 7 amplifiers A6, A7 areconnected in series across the load, and hence each amplifier will havean effective load impedance of one-half the impedance of loudspeaker S.Therefore if the latter is of the high impedance type, such as sixteenohms, each amplifier will have the better matched effective load ofeight ohms.

It will thus be seen that the consumer has his choice of either twolow-power stereo channels, or a single high-power monophonic amplifierwith an impedance-matching capability for increased power output. Thismodular arrangement enables the consumer to start his stereo system at aminimum expense by purchasing a pair of channels and utilizing thestereo mode connection, and then to increase the power capability of thesystem by purchasing another pair of channels and utilizing each pair ineither the parallel or series mode as may be suitable for the speakerimpedances. In this way the system is improved without sacrificing theequipment originally purchased from consideration of modest initialcost.

Referring now to FIG. 8, the circuit embodiment there disclosedcomprises an emitter-follower input stage including an NPN transistorQ1a having its base coupled through capacitor C1a to the hot inputterminal I1a. The other input terminal 12a is grounded as shown. Theemitter of transistor Q1a is provided with a load resistor R3a extendingto ground.

There is provided a fixed regulated potential source at the negativeelectrode of a zener diode Za having its positive electrode connected toground A. A resistor R4a extends from the negative terminal B₁ - of aconventional non-regulated power supply (not shown) to the negativeelectrode of zener diode Za. In order to filter out any residual ripple,a capacitor C3a may be connected in parallel across zener diode Za. Thebase of transistor Q1a is biased by connection to the junction of a pairof resistors R1a, R2a connected in series beween the negative electrodeof zener diode Za and ground. The collector of transistor Q1a may alsobe connected to said negative electrode of zener diode Za.

The second stage including PNP transistor Q2a is the first of the twocommon-emitter stages. The base of transistor Q2a is coupled bycapacitor C2a to the emitter of transistor Q1a and is biased byconnection to the junction of the series-connected pair of resistors R5aand Pa, the latter preferably being variable in the form of apotentiometer for adjustment of the quiescent direct-current level ofthe output stage in a manner to be described. The negative end ofpotentiometer Pa is connected to the junction of resistor R4a and zenerdiode Za, and the positive end of resistor R5a is connected to ground. Acollector load resistor R9a extends from the collector of transistor Q2ato the negative supply terminal B₁ -.

The second common-emitter stage comprises an NPN transistor Q3a havingits base direct-current coupled to the collector of the firstcommon-emitter transistor Q2a. The emitter of transistor Q3a isconnected through bias resistor R12a to A.C. ground at the negativesupply terminal B₁ - and a conventional bypass capacitor C7a isconnected in parallel across resistor R12a. The collector load impedanceof transistor Q3a comprises the series connection of resistors R10a,R11a and temperature-compensating bias diodes D1a, D2a. The upper end ofresistor R10a is connected to the positive power supply terminal B+ andthe lower end of diode D2a is connected to the collector of transistorQ3a. If warranted by the phase-shift characteristics of the amplifier, acapacitor C5a may be connected between the collector and base oftransistor Q3a to improve the feedback stability margin at highfrequencies. A conventional bootstrapping capacitor C6a extends fromoutput bus Oa to the junction of resistors R10a, R11a.

Connected in series between the emitter of transistor Q2a and the outputbus Oa are a pair of resistors R7a, R8a. The junction of the latter isgrounded with respect to alternating-current signals by a capacitor C4a.A resistor R6a is connected at one end to the output bus Oa and at theother end to the emitter of transistor Q2a. Resistors R7a, R8a thusprovide direct-current feedback from the output to the emitter oftransistor Q2a to maintain the quiescent direct-current potential ofoutput terminal O1a at ground level and thereby obviate any substantialdirect-current through the voice-coil of speaker S. At signalfrequencies alternating current feedback is provided by resistors R6a,R7a.

Transistors Q2a and Q3a are of opposite polarity types, the former beingshown for purposes of illustration as of type PNP whereas the latter isshown as type NPN. This enables the base of transistor Q3a to bedirect-current-coupled to the collector of transistor Q2a while theemitter of the latter may be direct-current-coupled through the feedbacknetwork R7a, R8a to the output O1a of the amplifier.

The drive stage comprises a pair of complementary transistors Q4a andQ5a, the former being shown as type NPN and the latter as type PNP. Thecollector of transistor Q4a is connected to the positive power supplyterminal B+ and its emitter is connected through bias resistor R13a tooutput bus Oa. The emitter of transistor Q5a is connected to output busOa by resistor R14a and its collector is connected by bias resistor R15ato the negative terminal B₁ -. The base of transistor Q4a isdirect-current-coupled to the junction of resistor R11a and diode D1a,and the base of transistor Q5a is direct-current-coupled to thecollector of transistor Q3a. Drive transistors Q4a, Q5a operate in apush-pull mode and are preferably biased for Class AB operation in theusual manner.

The output stage comprises a pair of output transistors Q8a and Q9aconnected in series between the positive power supply terminal B+ andthe negative terminal B₁ -, in the conventional single-ended push-pullconfiguration. The collector of transistor Q8a is connected to thepositive supply terminal B+ and its emitter is connected throughresistor R20a to the output bus Oa. Also connected to the latter is thecollector of transitor Q9a having its emitter connected through resistorR21a to the negative supply terminal B₁ -. The base of transistor Q8a isdirect-current-coupled to the emitter of NPN drive transistor Q4a andthe base of output transistor Q9a is similarly direct-current-coupled tothe collector of PNP drive transistor Q5a. The resulting combination ofdrive stage Q4a, Q5a and output stage Q8a, Q9a constitutes thewell-known quasi-complementary-symmetry arrangement.

A load, such as illustrated by loudspeaker S, is direct-coupled betweenthe hot output terminal O1a and the grounded output terminal O2a. Itwill thus be seen that if the quiescent direct-current potential ofoutput terminal O1a departs substantially from ground level a directcurrent will flow through the voice-coil of speaker S. This current willbias the cone of speaker S to an off-center position so as to increasethe harmonic and intermodulation distortion characteristics of thespeaker. Therefore it is important that the quiescent direct-currentpotential of output terminal O1a be maintained substantially at groundlevel.

This is achieved by a large amount of direct-current feedback from theoutput terminal O1a through the feedback network R7a, R8a to the emitterof transistor Q2a which is made possible by the existence of two stagesof gain within the feedback loop; that is, the overall forwardtransmission gain is equal to the product of the individual voltagegains of the two common-emitter stages Q2a and Q3a. This large amount ofdirect-current feedback does not affect the stability characteristics atsignal frequencies because at such frequencies the direct-currentfeedback is bypassed to ground by capacitor C4a and the amount ofalternating-current feedback may be selected as desired by choosing theproper magnitude ratio of resistor R6a to resistor R7a.

Due to economic considerations, the respective potentials at powersupply terminals B+ and B₁ - are unregulated, so as to undergo asubstantial voltage drop and to contain a high ripple content whensubstantial power is drawn from the supply. Injection of the powersupply ripple into the first two stages Q1a, Q2a is prevented by thebias network including zener diode Za, thereby avoiding theamplification of the ripple and maintaining a low hum level at theoutput of the amplifier.

Zener diode Za also maintains the base of transistor Q2a at a relativelyfixed direct-current potential independent of variations in the powersupply voltage due to regulation under load or voltage variations in thesupply mains. This is important because the direct-current feedbacknetwork is arranged so that the potential at output terminal O1a iscompared with that at the base of transistor Q2a which is therefore thereference potential. The feedback network corrects for any deviation ofthe potential of output terminal O1a from this reference potential andif the latter is substantially fixed the potential of output terminalO1a may be maintained at the proper ground level so as to preventquiescent direct-current through the voice-coil of speaker S.

Another advantage of the circuit of FIG. 8 is that the distortion may bereduced to extremely small levels by the application of a large amountof alternating-current feedback without the usual problems ofoscillation and poor transient response to an inadequate stabilitymargin. With respect to low frequencies, unstable operation such as"motor boating," blocking bias changes, and other symptoms ofinstability cannot occur because the circuit has no low-frequencyphase-shift-producing coupling capacitors within the feedback loop.

With respect to high frequencies, the circuit arrangement providesstability by causing the second common-emitter transistor Q3a to rolloff at a much lower frequency than the other stages so that the overallamplifier gain is reduced to unity before the total phase shift reaches180°. This is achieved by driving transistor Q3a from a relativelyhigh-impedance source provided by the preceding first common-emitterstage Q2a. The impedance of the driving source is substantially thevalue of resistor R9a which is preferably about several thousand ohms.Hence the second common-emitter stage Q3a will roll off substantially atthe beta cutoff frequency. On the other hand, the first common-emitterstage Q2a is driven by the relatively low-impedance source provided bythe preceding emitter-follower transistor Q1a. Hence transistor Q2a hasan effective high-frequency cutoff substantially beyond the beta cutofffrequency. The drive-output stage configuration comprising transistorsQ4a, Q5a and Q8a, Q9a also operates in the emitter-follower mode, sothat the voltage response of this configuration may also extend farbeyond the beta cutoff frequency of the second-common-emitter stage Q3a..Iadd.The above-described combination of low source impedance for thefirst common-emitter stage and high source impedance for the secondcommon-emitter stage serves to enhance the inherent mode of operation ofthe disclosed cascade arrangement of two successive common-emitterstages followed by an emitter-follower stage. That is, the firstcommon-emitter stage has a relatively high roll-off frequency because itis loaded by the inherently low input impedance of the secondcommon-emitter stage, and the second common-emitter stage has arelatively low roll-off frequency because it is loaded by the inherentlyhigh input impedance of the emitter-follower operation of thedrive-output stages.

This mode of operation is also inherent in the other disclosedembodiments comprising the cascade arrangement of two successivecommon-emitter stages followed by an emitter-follower drive-outputconfiguration. However, the circuits of FIGS. 2, 4, 8 and 9 areparticularly advantageous in that this mode of operation is achievedalong with the important capability of using output transistors of thesame polarity type. .Iaddend.As a result, alternating-current feedbackresistors R6a, R7a may be selected so as to apply an unusually largeamount of feedback without causing oscillation, transient ringing, orother symptoms of insufficient stability margin at the high-frequencyend of the spectrum.

It will be seen that when drive transistor Q4a is cutoff, no currentflows through bias resistor R13a and hence the base of output transistorQ8a is at the same potential as output bus Oa. Hence in the absence ofthe circuitry to be described below the base and emitter of transistorQ8a would be substantially at the same potential and no reverse biaswould be applied to the base-emitter junction of output transistor Q8aduring its "off" half of the cycle. Similarly, when drive transistor Q5ais cutoff, no current flows through bias resistor R15a and hence thebase and emitter of output transistor Q9a would be substantially at thesame potential so that no reverse bias would be applied to thebase-emitter junction of transistor Q9a during its "off" portion of thecycle.

As a result, at high frequencies excess minority carriers would remainstored in the respective bases of output transistors Q8a, Q9a duringtheir "off" portions of the cycle in the absence of any reverse bias todrain off the trapped majority carriers which keep an equal number ofminority carriers in the base in accordance with the requirement forspace-charge neutrality. A substantial collector current would therebyflow through each output transistor Q8a, Q9a when it should be cutoff,at which time the transistor is subjected to a relatively high collectorvoltage. This would cause excessive heat dissipation and destruction ofoutput transistors Q8a, Q9a when subjected to a continuous high-levelsignal at a high audio frequency.

To obviate this problem, the circuit of FIG. 8 is provided with a pairof transistors Q8a and Q7a which function to provide a reverse bias atthe respective base-emitter junctions of output transistors Q8a, Q9aduring their respective "off" portions of the push-pull cycle. Thecollector of transistor Q6a is connected to the base of outputtransistor Q8a and the base of transistor Q6a is connected to thecollector of driver transistor Q5a. The emitter of transistor Q6a isconnected through bias resistor R16a to the negative supply terminal B₁-. The collector of transistor Q7a is connected to the base oftransistor Q9a. The base of transistor Q7a is connected through resistorR17a to the output bus Oa and also through resistor R18a to an auxiliarynegative supply terminal B₂ -. Also connected to the latter through abias resistor R19a is the emitter of transistor Q7a.

During the "off" phase of output transistor Q8a, a substantial currentis drawn through bias resistor R15a by drive transistor Q5a so as toraise the potential at the base of transistor Q6a thereby turning thelatter "on" to draw current from output bus Oa through resistor R13a tothe collector of transistor Q6a. As a result the potential at the baseof output transistor Q8a will be below that of output bus Oa by theamount of the potential drop through resistor R13a, therebyreverse-biasing the base-emitter junction of output transistor Q8a.Similarly, during the "off" phase of output transistor Q9a the potentialof output bus Oa rises so as to raise the potential at the base oftransistor O7a thereby turning the latter "on" so as to draw currentfrom the negative supply terminal B₁ - through the bias resistor R15a tothe collector of transistor Q7a thereby reverse-biasing the base-emitterjunction of transistor Q9a by approximately the amount of the voltagedrop in resistor R15a.

Referring now to FIG. 9, there is shown a modified form of the inventionwherein the base of the first common-emitter stage is biased by directcoupling to the emitter of a preceding emitter-follower stage. The inputterminal I1b is coupled by a capacitor C1b and a resistor R1b to thebase of an NPN transistor Q1b operating in the emitter-follower mode.The base of transistor O1b is biased by a variable resistance in theform of a potentiometer Pb connected at one end to ground and at theother end in series with a resistor R2b extending to said base. Theother input terminal I2b is grounded as shown. The collector oftransistor Q1b is connected to ground and its emitter is connectedthrough emitter load resistor R3b and filter resistor R17b to thenegative supply terminal B- of a conventional non-regulated power supply(not shown). A capacitor C5b extends from the junction of resistors R3b,R17b to ground to bypass the ripple at supply terminal B-.

The emitter of transistor Q1b is direct-coupled to the base of a PNPtransisor Q2b constituting the first common-emitter stage. The emitterof transistor Q2b is connected through the series combination ofresistor R4b and capacitor C2b to ground, and its collector is connectedto the filtered negative supply through the load resistor R5b. Thecollector of transistor Q2b is direct-coupled to the base of an NPNtransistor Q3b constituting the second common-emitter stage and havingits emitter connected to the filtered negative supply through biasresistor R9b in parallel with bypass capacitor C4b. The collector loadimpedance of transistor Q3b comprises a series network including loadresistors R7b, R8b and temperature-compensating diodes D1b, D2bextending from the positive supply terminal B+ to said collector. Aconventional bootstrapping capacitor C3b has one end connected to theoutput bus Ob extending from the hot output terminal O1b and its otherend connected to the junction of resistors R7b, R8b.

It will be noted that each transistor of the cascade arrangement Q1b,Q2b, Qb3 is of complementary polarity type with respect to the precedingtransistor. For purposes of illustration, transistors Q1b and Q3b areshown to be of NPN type whereas transistor Q2b is illustrated as of PNPtype. However, it will be understood that these polarity types may bereversed if desired. As a result of this arrangement direct-couplingbetween these stages with maintenance of proper bias conditions can beobtained without the use of Zener diodes, regulated power supplies andsimilar expedients.

The drive stage comprises a pair of complementary transistors Q4b andQ5b, for purposes of illustration the former being of NPN type and thelatter of PNP type. The collector of transistor Q4b is connected to thepositive supply terminal B+ and its emitter is connected through biasresistor R10b to output bus Ob. Also connected to the latter throughresistor R11b is the emitter of transistor O5b having its collectorconnected through bias resistor R12b to the negative supply terminal B-.The base of drive transistor Q4b is direct-coupled to the junction ofresistor Q8b and diode D1b, and the base of drive transistor Q5b issimilarly direct-coupled to the junction of diode D2b and the collectorof transistor Q3b.

The next stage has a novel function and will be referred to by thecoined term "quasi-output" stage. It comprises a pair of transistorsQ6b, Q7b of the same polarity type. The collector of transistor Q6b isconnected to the positive supply terminal B+ and its emitter isconnected to output bus Ob through resistor R13b. Also connected tooutput bus Ob is the collector of transistor Q7b having its emitterconnected through resistor R14b to the negative supply terminal B-. Thebase of quasi-output transistor Q6b is direct-coupled to the emitter ofdrive transistor Q4b and the base of the other quasi-output transistorQ7b is similarly direct-coupled to the collector of drive transistorQ5b.

The output stage comprises a pair of transistors Q8b, Q9b of the samepolarity type. The collector of transistor Q8b is connected to thepositive supply terminal B+ and its emitter is connected throughresistor R15b to output bus Ob. Also connected to the latter is thecollector of output transistor Q9b having its emitter connected throughbias resistor R16b to the negative supply terminal B-. The base ofoutput transistor Q8b is direct-coupled to the emitter of quasi-outputtransistor Q6b and the base of output transistor Q9b is direct-coupledto the emitter of quasi-output transistor Q7b.

The reference letter S indicates a loudspeaker or other load having oneend connected to the hot output terminal O1b and its other end connectedto the second output terminal O2b which may be grounded as shown. Inorder to prevent the flow of quiescent direct-current through thevoice-coil of speaker S, the quiescent potential of output terminal O1bmust be maintained substantially at ground level. The bias conditions ofthe various stages may be initially set so as to adjust the potentialoutput terminal O1b to ground level by varying potentiometer Pb.

Since the base of transistor Q1b is connected to ground through resistorR2b and potentiometer Pb, the base of transistor Q1b is maintained at asubstantially fixed potential with respect to ground. Because transistorQ1b operates in the emitter-follower mode, its emitter will bemaintained at a substantially constant predetermined potentialdifference from that of its base independently of any potentialvariation and ripple component at the negative power supply terminal B-.As a result the base of the first common-emitter transistor Q2b ismaintained at a relatively fixed reference potential.

The potential at the emitter of transistor Q2b is constantly comparedwith this fixed reference potential at its base by means of thedirect-current feedback network comprising feedback resistor R6bextending from output bus Ob to the emitter of transistor Q2b. Forexample, if the potential of output bus Ob tends to rise above groundlevel the resulting increased voltage across the base-emitter junctionof transistor Q2b causes more collector current to flow therethroughthereby increasing the voltage drop across resistor R5b and increasingthe potential at the collector of transistor Q2b. This in turn raisesthe potential at the base of transistor Q3b so as to increase thecollector current therethrough and thereby increase the voltage dropacross load resistors R7b, R8b. This lowers the potential at thecollector of transistor Q3b. Since the remaining drive, quasi-output andoutput stages operate effectively in the emitter-follower mode, thelower potential of the collector of transistor Q3b in turn results in alower potential of output bus Ob so as to counteract the originallyassumed tendency of the potential of output bus Ob to rise. It will beobvious that the feedback arrangement has the opposite effect in theevent that there is a tendency for the potential of output bus Ob tofall. The unity feedback provided by the resistor R6b is reduced atsignal frequencies by capacitor C2b and resistor R4b. The relativemagnitudes of resistors R4b, R6b are selected so as to attain thedesired amount of alternating-current feedback.

The amount of this alternating-current feedback may be made extremelylarge so as to reduce the distortion of the amplifier to almostunmeasurable levels while retaining an ample stability margin which willobviate any tendency to oscillation, ringing, or transient distortion,as discussed above with respect to the circuit modification of FIG. 8.At the high frequency end the circuit of FIG. 9 provides an additionalstability margin by virtue of the fact that the lower output transistorQ9b is driven by a source of much lower impedance than that which drivesthe lower output transistor Q9a in FIG. 8. That is, the magnitude ofresistor R15a of FIG. 8 is preferably about one hundred ohms which isthe source of impedance seen at the base of output transistor Q9a. Thiscauses output transistor Q9a to roll-off at substantially its betacutoff frequency. In the present state of the art, this beta cutofffrequency for output transistors of reasonable cost is approximately oneoctave below the upper audio limit, generally .[.regard.]..Iadd.regarded .Iaddend.as 20 kHz. The resulting phase shift in thecircuit of FIG. 8 substantially reduces the stability margin at highfrequencies.

In the circuit of FIG. 9, on the other hand, the lower output transistorQ9b is driven by a relatively low source impedance of only a few ohms.As will be explained below, the magnitude of resistor R14b is preferablyin the range of about 2 to 10 ohms. Furthermore, the base of transistorQ9b is driven by the emitter of quasi-output transistor Q7b operating inthe emitter-follower mode which provides a low source of impedance. As aresult of the low impedance of the driving source seen by the base ofoutput transistor Q9b the effective roll-off frequency of the latterextends far beyond the beta cutoff frequency of the transistor and theresulting reduction of phase shift improves the stability margin of thecircuit of FIG. 9 as compared to that of FIG. 8.

In FIG. 9 the quasi-output stage Q6b, Q7b and output stage Q8b, Q9bfunction in the novel manner disclosed in my prior copending applicationentitled, "Transistor Audio Amplifier with Power-Division OutputStages," Ser. No. 501,515, filed Oct. 22, 1965. The quasi-output stageis normally biased with a small quiescent current for Class AB push-pulloperation so as to supply power to the load S at low signal levels whileat these levels the output stage Q8b, Q9b is entirely cutoff. The loadcurrent flows alternately through resistors R13b, R14b which arepreferably in the range of about 2 to 10 ohms each. At higher signallevels the respective voltages across R13b, R14b attain a magnitudesufficient to turn on respective output transistors Q8b, Q9b so that thelatter then become active and supply power to the load S at highersignal levels, at which time quasi-output transistors Q6b, Q7b functionas drive transistors for the output stage Q8b, Q9b.

Since there is substantially no quiescent bias current in the outputstage Q8b, Q9b even at higher operating junction temperatures, theproblems of thermal runaway, excessive power dissipation and reducedpower ratings are avoided. Because the low-level power is provided bythe quasi-output stage Q6b, Q7b which is biased with a sufficientlylarge quiescent current for this purpose, crossover distortion at lowlevels is minimized. No problem of excessive bias current can arise inquasi-output stage Q6b, Q7b because this stage provides only smallamounts of power and therefore operates at relatively cool junctiontemperatures so that the transfer characteristic of the transistors doesnot vary significantly. For further details as to the mode of operationand advantages of this drive-output arrangement reference is made tosaid prior copending application Ser. No. 501,515.

Alternatively, transistors Q6b, Q7b of FIG. 9 may function as aconventional emitter-follower drive stage and the output stage Q8b, Q9bmay be provided with a small quiescent bias current so as to operate inthe Class AB mode in the usual manner. In this event emitter resistorsR13b, R14b are of a magnitude sufficient to bias output transistors Q8b,Q9b at this quiescent point, a typical magnitude for these resistorsbeing about two hundred ohms. In this arrangement the lower outputtransistor Q9b will still be driven by a relatively low source impedancesince transistor Q7b operates in an emitter-follower mode having a lowoutput impedance. As a result the extended high-frequency response andreduced phase shift of transistor Q9b discussed above are retained.

In all of the embodiments disclosed in FIGS. 1 to 9 described above andin FIGS. 1a, 2a, 3a and 4a to be described below the critical importanceof the drive stage for the practical utilization of the subject circuitsin most applications, and the coaction of the drive stage with thedirect-current coupling arrangement for this purpose, will not beapparent to those skilled in the art and are now described. Althoughproviding obvious secondary advantages such as increased current gainand, in some embodiments, phase inversion, the primary need for thedrive stage is to protect the loudspeaker from being damaged in theevent of failure of an output transistor. This requirement arisesbecause of the lack of an output coupling capacitor or outputtransformer to isolate the loudspeaker from the power supply in theevent of a shorted output transistor. The drive stage is able to providethe required protection of the loudspeaker because of the direct-currenttransmission path through the amplifier. Furthermore, the protectiveeffect is greatly enhanced by the direct-current feedback network, asexplained in detail below.

The importance of this safety feature cannot be under-estimated. Theexpense and inconvenience of delivering a twenty-pound amplifier to arepair shop for replacement of a few transistors is far less than thatinvolved in returning a one-hundred-pound speaker system to the factoryfor replacement of the voice-coil, cone and suspension. Furthermore,although the consumer has learned to accept the risk of transistoramplifier failure as inherent in the price he must pay for itsadvantages over tube amplifiers, reputable manufacturers areunderstandably reluctant to market an amplifier which might also destroythe loudspeaker.

The operation of the subject protective arrangement will be describedwith respect to FIG. 1, it being understood that the same protectiveoperation is provided by the other embodiments of FIGS. 2 to 4, 8 and 9and that a substantially similar operation is provided by theembodiments of FIGS. 1a, 2a, 3a and 4a to be described below. Referringto FIG. 1, let it be assumed that output transistor Q5 undergoes ashort-circuit failure due to second breakdown or other cause. In thisevent output terminal O1 will be connected practically directly to thenegative terminal of the power supply B'1 since the magnitude of emitterresistor R12 is generally only about one-half ohm. Almost the entirevoltage of power supply B'1 is thereby impressed across the voice-coilof loudspeaker S so that in the absence of some protective circuitarrangement a large surge of current will flow through the voice-coil soas to burn out the voice-coil or to cause mechanical damage to thevoice-coil or the cone suspension when the cone "buttoms." Theprotective arrangement of the present invention prevents a large currentsurge through the voice-coil, in the following manner.

Since the bases of drive stage transistors Q2, Q3 are direct-currentcoupled by leads 11, 12 to the collector circuit of transistor Q1 saidbases are maintained at the respective potentials determined by thesignal applied to the input terminal I1 of the amplifier notwithstandingthe short-circuit failure of output transistor Q5. The tendency of saidbases to resist potential deviation due to output transistor failure isfurther enhanced by the direct-current feedback network comprising lead16, potentiometer Pb and resistor R2 so as to apply to the base oftransistor Q1 a feedback signal counteracting any tendency of thepotentials of the drive stage bases to go negative. Therefore as thepotential of output terminal O1 commences to go negative in response tothe short-circuit failure of output transistor Q5 the emitter of drivetransistor Q2 goes negative therewith to increase the forward-biasvoltage across the base-emitter junction of drive transistor Q2. Thelatter thereby conducts a large collector current. The resulting largevoltage drop across resistor R8 causes a large forward-bias across thebase-emitter junction of lower output transistor Q4 to cause the latterto conduct a large collector current.

The currents thus conducted by transistors Q2 and Q4 flow from thepositive terminal of power supply B1 through said transistors to outputterminal O1, through emitter resistor R12, and then through theshort-circuited output transistor Q5 to the negative terminal of powersupply B'1. As a result a large portion of the short-circuit currentflowing through output transistor Q5 is supplied by transistors Q2 andQ4 so as to be effectively bypassed around loudspeaker S instead offlowing therethrough. The voice-coil of loudspeaker S is therebysubjected to a substantially smaller surge of current so as to obviatethe damage that would otherwise occur.

The usual drive transistor types employed will act as protective fusesby undergoing almost instantaneous second breakdown in response tobreakdown of the output transistor in the opposite half of the push-pullarrangement. That is, instead of merely passing a large collectorcurrent, transistor Q2 will instantaneously fail and become ashort-circuit thereby more effectively bypassing the short-circuitcurrent around loudspeaker S and also serving to protect the loweroutput transistor Q4 to prevent the latter from undergoing breakdown.

If output transistor Q4 should break down and undergo a short-circuitfailure, instead of output transistor Q5, then the other drive stagetransistor Q3 will provide current bypass protection for loudspeaker Sin a similar manner. The direct-current coupling to the base oftransistor Q3 and the direct-current feedback network to the base oftransistor Q1 tend to maintain the potential of the base of transistorQ3 against displacement as the emitter potential of transistor Q3 isdrawn towards the positive potential of power supply B1 due to theshort-circuit of output transistor Q4. The forward-bias voltage acrossthe base-emitter junction of transistor Q3 is thereby increased and alarge emitter current flows through transistor Q3 to drive outputtransistor Q5 heavily into the active region. A large portion of theshort-circuit current flowing through transistor Q4 is thus supplied bytransistors Q3 and Q5 so as to reduce substantially the surge of currentthrough loudspeaker S. If drive transistor Q3 is of the type usuallyselected for this function it is likely to undergo almost instantaneoussecond breakdown so as to improve the efficacy of this protectivearrangement.

It will be understood that drive transistors Q2 and Q3, although in asense operate as fuses particularly when they undergo second breakdown,provide a protective function which could not be rendered even by thefastest acting instrument fuses available. It is well recognized thatfuses are ineffective to protect the loudspeaker because the latter isusually damaged before the finite time required for the fuse to melt.However, the time required for drive transistors Q2 and Q3 to undergosecond breakdown is so small as to result in practically instantaneousprotection for the loudspeaker.

In the embodiments of FIGS. 1 to 4, 8 and 9 the protective function ofthe drive stage is achieved by conductively coupling the drivetransistor emitters to the output terminal through bias resistors,whereas in the embodiments of FIGS. 1a, 2a, 3a and 4a the drive stageemitters are conductively coupled to the output terminal through thebase-emitter junctions of the respective output transistors, as will bedescribed below.

Referring now to FIG. 1a there is disclosed another embodiment of theinvention utilizing the same principles described above with respect toFIGS. 1 to 4. Input terminal I1' is coupled by capacitor C1' to the baseof PNP transistor Q1' operating Class A in the common-emitter mode. Theother input terminal I2' is grounded. In this embodiment the ground,instead of a zener diode, serves as a source of reference potential inthat bias resistor R1' extends from ground to the base of transistor Q1'to maintain the base at a fixed quiescent reference potentialsubstantially independent of power supply regulation and ripple, ambientand operating temperature variations, and other variables.

A resistor R2' and capacitor C2' are connected in series between theemitter of transistor Q1' and ground. Also connected to the emitter oftransistor Q1' is one end of a resistor R5' having its other endconnected to output bus O' of the amplifier. A fixed resistor R6' and avariable resistor in the form of a potentiometer P' are connected inseries between the B+ power supply terminal and output bus O'. Aresistor R6" extends from output bus O' to the negative terminal B- ofthe power supply. A resistor R4' extends between the junction ofresistor R6' and potentiometer P' and the junction of capacitor C2' andresistor R2'.

The function of resistors R4', R6' and potentiometer P' is to bias theemitter of transistor Q1' at the quiescent potential which is positivewith respect to ground. This is necessary because of the voltage dropsacross the base-emitter junction of transistor Q1' and the bias resistorR1'. It will be obvious that if the quiescent potential of the base isdisplaced in a predetermined direction and magnitude with respect toground, as compared with the above-described embodiments, then thequiescent potential of the emitter must be identically displaced inorder to maintain the same bias voltage across the base-emitter junctionof the transistor.

The second stage comprises an NPN transistor Q2' operating Class A inthe common-emitter mode and having its base direct-coupled to thecollector of transistor Q1', its emitter A.C.-coupled to ground throughthe negative supply terminal B-, and its collector connected to thelower end of a first load resistor R8' having its upper end connected toa second load resistor R7' in turn connected to the positive supplyterminal B+. A capacitor C4' may be provided between the collector andbase of transistor Q2' to provide a roll-off and phase shift at highfrequencies so as to increase the stability margin of the amplifier. Theusual bootstrapping capacitor C3' extends from output bus O' to thejunction of load resistors R7', R8'.

The third stage comprises a PNP transistor Q3' having its basedirect-coupled to the collector of transistor Q2', its emitterconductively connected to output bus O' and its collector connectedthrough load resistor R9' to the negative supply terminal B-. Theconnection of the emitter to output bus O' provides so-called"unity-gain feedback" for a purpose to be described.

The fourth stage comprises an NPN transistor Q4' operating Class A inthe common-emitter mode and having its base direct-coupled to thecollector of transistor Q3', its emitter connected to the negativesupply terminal B-, and its collector connected to the lower end of theusual temperature-compensating bias diodes W1', W2'. Connected in serieswith the latter is a bias resistor R12' and a pair of collector loadresistors R10', R11'. The upper end of resistor R10' is connected to thepositive supply terminal B+. Here again a phase shift capacitor C6' maybe provided between the collector and base of transistor Q4' and theusual bootstrapping capacitor C5' extends between output bus O' and thejunction of collector load resistors R10', R11'.

The complementary push-pull drive stage comprises NPN transistor Q5' andPNP transistor Q6' having their emitters connected in series by a biasresistor R13'. The collector of transistor Q5' is connected to thepositive supply terminal B+ and the collector of transistor Q6' issimilarly connected to the negative supply terminal B-. The base oftransistor Q5' is direct-current coupled to the lower end of collectorload resistor R11' and the base of transistor Q6' is direct-coupled tothe collector of transistor Q4'.

The push-pull output stage, unlike that of the previous embodimentsdescribed above, is of the complementary type and comprises an NPNoutput transistor Q7' and a PNP output transistor Q8' having theiremitters connected to output bus O'. The base of transistor output baseQ7' is direct-coupled to the emitter of drive transistor Q5' and thebase of output transistor Q8' is similarly direct-coupled to the emitterof drive transistor Q6'. The drive transistor emitters are thusconductively connected to output bus O' through the respectivebase-emitter junctions of output transistors Q7', Q8' so as to protectloudspeaker S from damage in the event of short-circuit failure of anoutput transistor.

For example, failure of output transistor Q7' will cause the potentialof output bus O' to swing upwardly toward the potential of supplyterminal B+ and will tend to pull the emitter of output transistor Q8'upwardly therewith to impart a large forward-bias to the base-emitterjunction of output transistor Q8'. The potentials of the base oftransistor Q8' and the emitter of transistor Q6' are thereby also drawnupwardly whereas the potential of the base of drive transistor Q6' ismaintained substantially fixed by the direct-current coupling theretothrough the previous stages and by the direct-current feedback network.Therefore the base-emitter junction of drive transistor Q6' becomesheavily foward-biased to provide a large drive current to the base ofoutput transistor Q8'. Both transistor Q6' and Q8' thus become highlyconductive so as to provide a substantial portion of the short-circuitcurrent flowing through output transistor Q7' and thereby substantiallyreducing the current surge through loudspeaker S. If drive transistorQ6' has the power capability of the types usually employed for thisfunction it is likely to go into second breakdown and fail so as toserve as an almost instantly acting protective fuse as described above.Since the drive and output stages of this embodiment are symmetrical, itwill be apparent that drive transistor Q5' functions in the same mannerin the event that output transistor Q8' breaks down and becomesshort-circuited.

Transistors Q9' and Q10' constitute a protective circuit to preventoutput transistors Q7' and Q8' from undergoing second breakdown in theevent of a short-circuit across output terminals O1', O2', or a.[.lead.]. .Iadd.load .Iaddend.impedance which is either highly reactiveor too low in magnitude, or any other condition which might cause theoperating point of one of the output transistors to enter a region ofhigh instantaneous current simultaneous with high instantaneous voltage.This might occur, for example, even with a normal speaker load if theamplifier is driven with a large low-frequency transient signal such asis generated when the tone arm is dropped upon the record or whenswitching between signal sources or when plugging and unpluggingconnector cables with the equipment energized. This protective circuitis substantially the same in principle and mode of operation to thatfirst disclosed in said prior application Ser. No. 388,399 except forminor differences arising from the fact that in said prior applicationthe current-sensing resistor was connected to the grounded outputterminal whereas in the present embodiment it is connected to the hotoutput terminal.

This current-sensing resistor is designated R17' in FIG. 1a and isconnected in series between output bus O' and the hot output terminalO1'. The respective emitters of transistors Q9', Q10' are connected toone end of resistor R17' and their respective bases are connectedthrough resistors R14', R15' to the opposite end of resistor R17'. Saidbases are also connected through resistors R16', R18' to ground. Thecollector of transistor Q9' is connected through diode D3' to the baseof transistor Q3', and the collector of transistor Q10' is similarlyconnected to said base through diode D4'.

As originally described in said prior application Ser. No. 388,399 theprotective circuit continuously senses both the instantaneous voltageand instantaneous current of the output transistors, and therebymonitors the instantaneous power dissipation of the output transistors.Should the operating point of either of the output transistors tend toenter a region of higher instantaneous power dissipation than thatpredetermined by the selected parameter values of the protectivecircuit, the latter will clip the drive signal from transistor Q2' totransistor Q3', in the following manner.

The load current passing through loudspeaker S from output stage Q7',Q8' must flow through resistor R17' so that the instantaneous voltageacross the latter is directly proportional to the emitter current of theoutput transistor which is conducting at that instant. Thiscurrent-responsive voltage is applied across the respective base-emitterjunctions of transistors Q9', Q10' through resistors R14', R15' tofoward-bias one junction and reverse-bias the other, depending upon thedirection of current flow through resistor R17'. For example, if thedirection is from output transistor Q7' to output terminal O1' then thebase-emitter junction of transistor Q9' will tend to be forward-biasedwhereas that of transistor Q10' will tend to be reverse-biased.

Opposing this bias is a signal applied to the respective bases oftransistors Q9', Q10' and which is proportional to the voltage swing ofthe midpoint of the output stage at the junction of the emitters oftransistors Q7', Q8'. For example, assuming that output transistor Q7'is conductive, the potential of the output stage midpoint will gopositive. Since resistor R17' is preferably in the range of aboutone-half ohm the potential of output terminal O1' will closely followthe rising potential at the midpoint of the output stage so as to raisethe potential at the emitter of transistor Q9'. However the base oftransistor Q9' does not rise in potential to the same extent because ofthe connection to ground through resistor R16' and hence the rise inpotential at output terminal O1' will tend to maintain transistor Q9'cutoff. Transistor Q9' will then have no effect upon the signaltransmitted by transistor Q4' to drive stage Q5', Q6'.

However, let it now be assumed that output terminals O1', O2' areinadvertently short-circuited and that the signal is rising in potentialso as to render output transistor Q7' conductive. Because of theshort-circuit, output terminal O1' remains at ground potential. Theemitter of output transistor Q7' can thus rise above ground potentialonly by the relatively small amount of the voltage drop across resistorR17', and hence the collector-to-emitter voltage of output transistorQ7' remains high. As the emitter current from transistor Q7' flowsthrough the short-circuit at the output terminal O1', O2' the potentialat the left-hand end of resistor R17' rises so as to forward-bias thebase-emitter junction of transistor Q9'. Resistor R16' does not in thisinstance provide a reverse-bias signal since the emitter of transistorQ9' remains at ground potential due to the short-circuit across outputterminals O1', O2'. Transistor Q9' thus becomes conductive toforward-bias diode D3' and draw current downwardly through collectorload resistors R7', R8'. The positively swinging signal at the base oftransistor Q3' is thereby instantaneously clipped so as to preventfurther drive through drive transistor Q5' to output transistor Q7'. Theoperating point of the latter is thereby prevented from rising towardthe region of high collector current and second breakdown.

It will be apparent that when output transistor Q8' is conductivetransistor Q10' will function in a similar manner in response to ashort-circuit across output terminals O1', O2' to clip the signalpotential at the base of transistor Q3' and thereby prevent theoperating point of output transistor Q8' from exceeding the secondbreakdown limit. It will be seen that the above-described mode ofoperation will also be effective when the load across output terminalsO1, O2 is of low magnitude so as to present an excessively steepload-line or is highly reactive such as may be due to a crossovernetwork or to a low-frequency transient.

The preferred parameters of the protective circuit components are asfollows: resistor R17':0.56 ohm; resistors R14', R15':100 ohms;resistors R16', R18':1,000 ohms.

The power supply comprises a transformer T having a primary winding T1connected through a slow-blow fuse F3 to the usual alternating-currenthouse line of about 117 volts applied at line terminals L1, L2. Thetransformer secondary winding T2 is provided with a center-tap groundedas shown. A conventional full-wave rectifier bridge comprisingrectifiers D5', D6', D7', D8' is connected to the transformer secondarywinding T2 and also to the filter capacitors C7' and C8'. The junctionof the latter is connected to ground at the negative terminal ofcapacitor C7' and the positive terminal of capacitor C8'. The positiveterminal of capacitor C7' constitutes the B+ terminal of the powersupply and the negative terminal of capacitor C8' constitutes the B-terminal of the power supply. Fast-blow instrument fuses F1, F2 are inthe B+ and B- supply lines respectively.

Output transistors Q7', Q8' are preferably biased in the cutoff regionso as to operate in the Class B mode. If both transistors are silicontypes then the base-emitter quiescent voltage of each transistor may beabout 300 millivolts which is substantially below the cut-in voltage.Therefore no quiescent current flows through output transistors Q7', Q8'except for a negligible leakage current, and hence there is no problemof thermal runaway.

Crossover distortion is obviated by the unity-gain feedback loop formedby the connection of output bus O' to the emitter of transistor Q3' andthe two stages of voltage gain provided by transistors Q3' and Q4'. Theportion of the amplifier following the collector of transistor Q2'operates effectively in the emitter-follower mode with consequent lowdistortion notwithstanding the cutoff Class B bias of the output stage.That is, any slight difference between the signal potential at the baseof transistor Q3' and that at output bus O' constitutes an error signalwhich is impressed across the base-emitter junction of transistor Q3'.This error signal is amplified by the latter and then further amplifiedagain by transistor Q4' so as to provide a doubly-amplified signal tothe drive and output stages with a polarity and magnitude to correct forthe error originally sensed at the base-emitter junction of transistorQ3'.

The complementary nature of the output stage is critical in thisarrangement. The large amount of feedback provided with the unity-gaininner feedback loop and the outer feedback loop is made feasible by thesymmetry of the complementary drive and output stages in thatphase-shift characteristics and correction networks suitable forone-half of these push-pull stages are also suitable for the other halfso that adequate high-frequency stability margins are obtainable forboth positive and negative portions of the signal swing. In the presentstate of the semiconductor art the use of a quasi-complementarydrive-output arrangement in combination with a unity-gain inner feedbackloop would inevitably result in "ringing," oscillation or othermanifestation of inadequate stability margin on at least either thepositive or negative portion of the signal.

The unity-gain feedback to the emitter of transistor Q3' is furtheradvantageous in that it extends the frequency response of the stageswithin the inner feedback loop, thereby improving the high-frequencystability margin of the outer feedback loop formed by the feedback tothe emitter of the first stage transistor Q1', in accordance with themode of operation described above with respect to the embodiments ofFIGS. 3 and 4.

In FIG. 1a there is shown a heavy dash-dot line enclosing a subnetworkcomprising the unity-gain feedback loop and the protective circuitincluding transistors Q9', Q10'. This subnetwork is designated by thereference letter A and is indicated symbolically in the below-describedembodiment of FIG. 4a to simplify the disclosure of the latter and toavoid repetition of the details of subnetwork A. To show connections tothe latter, five nodes thereof are identified in FIG. 1a by thereference designations N1 to N5 inclusive.

These nodal reference designations are also used in FIGS. 2a and 3a toshow the manner in which modified forms of the first two stages areconnected to the subnetwork A at the respective nodes. Referring firstto FIG. 2a, hot input terminal I3 is coupled by capacitor C9 andresistor R19 to the base of a PNP transistor Q11 operating Class A inthe common-emitter mode. The source of reference potential is at groundlevel and a bias resistor R20 extends from ground to the base oftransistor Q11. A load resistor R21' extends from the collector oftransistor Q11 to node N5. The other input terminal I4 is connected toground.

Alternating-current and direct-current feedback, as well as a quiescentbias voltage, are applied to the emitter of transistor Q11 by thefollowing circuit arrangement. A capacitor C10 and resistor R21 extendin series from ground to the emitter of transistor Q11. Also connectedto said emitter is one end of a feedback resistor R22 having its otherend connected to the positive terminal of a diode D9 the negativeterminal of which is connected to node N2 which in turn is connected tooutput bus O' of subnetwork A. A resistor R23 extends from the positiveterminal diode D9 to node N1 in turn connected to the B+ supplyterminal, and another resistor R24 extends from node N2 to node N5 inturn connected to the B- supply terminal.

The collector of transistor Q11 is direct-coupled to the base of an NPNtransistor Q12 also operating Class A in the common-emitter mode. Theemitter of transistor Q12 is connected to node N5 and its collector isconnected to node N3 in turn connected to the base of transistor Q3' ofsubnetwork A. A high-frequency phase shift capacitor C11 may be providedfrom the collector to the base of transistor Q12. A pair of loadresistors R22', R23' extend in series from the collector of transistorQ12 to node N1. A bootstrapping capacitor C11' extends from mode N2 tothe junction of resistors R22', R23'.

It will be seen that resistor R23 and diode D9 constitute a voltagedivider to maintain the potential at the right-hand end of resistor R22at a predetermined constant level above the potential of output bus O'.Hence the quiescent potential of the latter may be at ground level andthe voltage drop across diode D9 compensates for the voltage dropsacross the base-emitter junction of transistor Q11 and across biasresistor R20.

Referring now to FIG. 3a, there is shown another modified form of thefirst two stages which may be direct-coupled in cascade with subnetworkA. The hot input terminal 15 is coupled by capacitor C12 and resistorR25' to the base of a PNP transistor Q13. Also connected to the base isone end of a bias resistor R25 having its opposite end connected toground to which is also connected the other input terminal I6. Since thesource of reference potential is the ground itself the voltage of thissource must obviously remain fixed with respect to ground irrespectiveof variations in the power supply, junction temperature, componentparameters or other changes.

The emitter of transistor Q13 is provided with alternating-current anddirect-current feedback signals and a quiescent bias potential in thefollowing manner. A resistor R26 has one end connected to the emitter oftransistor Q13 and its other end connected to a capacitor C14 in turnconnected to ground. A resistor R28 extends from node N1 to one end of avariable resistor in the form of a potentiometer P having its other endconnected to node N2. A bypass capacitor C13 is connected in parallelacross potentiometer P. A feedback resistor R27 extends from the lowerend of resistor R28 to the emitter of transistor Q13. A resistor R29extends between node N2 and node N5.

Resistor R28 and potentiometer P function as a voltage divider tomaintain the potential of the right-hand end of resistor R27 at aconstant level above the potential of output bus O' of subnetwork A.Should the quiescent potential of output bus O' tend to drift withrespect to ground so as to provide a direct-current offset, thepotential at the right-hand end of resistor R27 will vary therewith soas to provide a direct-current feedback signal to the emitter oftransistor Q13 with a magnitude and of a polarity to counteract thistendency.

The collector of transistor Q13 is provided with a load resistor R26'extending to node N5 in turn connected to the B- supply terminal. Thebase of the second stage transistor Q14 is direct-coupled to thecollector of transistor Q13. A pair of load resistors R30, R31 extend inseries from the collector of transistor Q14 to node N1 which is in turnconnected to the B+ supply terminal. A bootstrapping capacitor C15extends from node N2 to the junction of resistors R30, R31. Aphase-shift capacitor C14' may be provided between the collector andbase of transistor Q14. The emitter of the latter is connected to nodeN5 in turn connected to the B- supply terminal. The collector oftransistor Q14 is direct-coupled to node N3 and therefore to the base oftransistor Q3' of subnetwork A.

In FIG. 4a there is shown another embodiment of the present inventionwherein each of the common-emitter stages consists of twoemitter-coupled transistors arranged substantially symmetrically so asto operate in the well-known differential amplifier node. Insofar as thedirect-current coupling aspect of the invention is concerned it isimmaterial as to whether each common-emitter stage is embodied in theform of a single grounded-emitter transistor or in the form of a pair ofemitter-coupled transistors since these forms are known equivalent andprovide the same mode of operation for the amplifier circuit as a whole.However, the symmetrical differential stages of FIG. 4a are advantageousin that the second common-emitter stage provides oppositely-phasedpush-pull signal, which may be utilized to drive the respective oppositehalves of a full-bridge output arrangement, as will be described below.

Referring to FIG. 4a in more detail, the hot input terminal I9 iscoupled by capacitor C24 and resistor R54 to the base of a PNPtransistor Q25. The other input terminal I10 is connected to ground fromwhich extends a bias resistor R53 to the junction of resistor R54 andcapacitor C24. The first common-emitter stage further comprises a secondPNP transistor Q26 having its base biased by a pair of resistors R62,R63 extending in series to ground. In the solid-line position of switchS1 resistor R62 is shorted. Resistor R63 is bypassed by a capacitor C25.

The emitters of transistors Q26', Q26 are coupled by a pair of biasresistors R57, R59 of relatively small magnitude and a resistor R56 ofrelatively large magnitude presents a substantially constant currentsource to the emitters. Resistor R56 extends from the B₁ + supplyterminal to the junction of resistors R57, R59. Load resistors R58, R60extend from the respective collectors of transistors Q25, Q26 to theB₁ - supply terminal. The split power supply is indicated schematicallyby the battery symbols designated PS₁ + and PS₁ - respectively. Thecenter-tap of this split power supply is grounded as shown.

The second common-emitter stage is also of the symmetricalemitter-coupled type operating in the differential amplifier mode but inthis stage the transistors Q27, Q28 are type NPN. The base of transistorQ27 is direct-coupled to the collector of transistor Q25 and the base oftransistor Q28 is direct-coupled to the collector of transistor Q26. Theemitters of transistors Q27, Q28 are coupled through bias resistors R65,R69 connected at a common junction from which extends a resistor R66 tothe B₁ - supply terminal. Load resistors R64, R68 extend from therespective collectors of transistors Q27, Q28 to the B₁ + supplyterminal.

The collector of transistor Q28 is direct-coupled to node N3 ofsubnetwork A indicated schematically in FIG. 4a and disclosed in detailin FIG. 1a. Node N1 of subnetwork A is connected to the B₁ + supplyterminal and node N5 is connected to the B₁ - supply terminal. Afeedback resistor R55 extends from node N2 to the base of transistor Q25to provide both direct-current and alternating-current feedback fromoutput bus O' of subnetwork A. Output terminal O1' of subnetwork A isconnected to the hot terminal S11 of loudspeaker S when switch S2 is inthe solid-line position shown. The other terminal ST2 of loudspeaker Sis grounded.

That portion of FIG. 4a thus far described constitutes a completeamplifier and operates in a manner similar to that shown in FIG. 1a. ThePNP common-emitter stage Q25, Q26 of FIG. 4a is the equivalent of PNPtransistor Q1' of FIG. 1a, and similarly, the NPN common-emitter stageQ27, Q28 of FIG. 4a is equivalent to transistor Q2' of FIG. 1a. In FIG.4a the feedback is applied through resistor R55 to the base oftransistor Q25 rather than to the emitter of the first stage transistoras in FIG. 1a because the signal at the collector of transistor Q28 isout-of-phase with that at input terminal I9. If desired, the output ofthe second common-emitter stage may be taken instead from the collectorof transistor Q27 which may be connected to node N3 to transmit to thelatter a signal in-phase with that at input terminal I9. In this eventthe feedback may be applied to the base of transistor Q26 which willthen transmit the feedback signal to the emitter of transistor Q25,thereby providing substantially the same feedback arrangement as thatshown in FIGS. 2, 4, 8, 9 and 1a. .Iadd.This arrangement is shown inFIG. 4b wherein the components are designated by the same referencenumerals as the corresponding components in FIG. 4a, followed by thesuffix "b". .Iaddend.

By virtue of the in-phase and out-of-phase signals at the collectors oftransistors Q27 and Q28 respectively, the embodiment of FIG. 4a permitsthe utilization of two single-ended half bridge drive-output circuits tobe connected either in a parallel mode similar to that shown in FIG. 6or in a full-bridge series mode similar to that shown in FIG. 7. Thesecond drive-output circuit is designated in FIG. 4a as A' and may beidentical to subnetwork A. Subnetwork A' together with its own splitpower supply PS₂ +, PS₂ - may be mounted on an auxiliary chassisseparate from the chassis containing subnetwork A and the common-emitterstages, and this auxiliary chassis may be marketed separately as anauxiliary power booster.

More specifically, node N1' of subnetwork A' is connected to the B₂ +supply terminal and node N5' is connected to the B₂ - supply terminal.When switch S3 is in the solid-line position shown the output terminalO1' of subnetwork A' is connected to speaker terminal ST1 so thatsubnetworks A and A' are connected in parallel with respect toloudspeaker S. This would be the preferred mode of operation ifloudspeaker S is of relatively low impedance, such as four ohms or less.In this parallel mode of operation switches S4 and S5 are in theposition shown in solid lines so as to connect nodes N2' and N4' ofsubnetwork A' to nodes N2 and N4 respectively of subnetwork A. Node N3'of subnetwork A' is similarly connected to node N3 of subnetwork A bythe solid-line position of switch S6.

For use with loudspeakers of eight ohms or higher impedance thefull-bridge series mode of operation may be preferred in order to obtaingreater power. The connections for this mode of operation are indicatedby the dashed-line positions of the respective switches which therebyconnect subnetworks A and A' in series with respect to loudspeaker S'shown in dash-dot lines. In this series mode of operation it will beseen that switches S2 and S3 connect the respective output terminals O1'of subnetworks A and A' to the speaker terminals ST3 and ST4 ofloudspeaker S'. Switch S6 connects the collector of transistor Q27 tonode N3' of subnetwork A' so that the latter is driven in opposite phaseto that of the signal derived from the collector of transistor Q28 whichdrives subnetwork A. Switches S1 and S4 connect node N2' throughfeedback resistor R61 to the base of transistor Q26 to provide anegative feedback signal to the latter. Switch S5 opens so that node N4'is no longer connected to node N4.

In this full-bridge mode of operation any tendency to direct-currentdrift of output terminals O1' from ground potential is reduced by thecommon-mode feedback provided by resistors R70, R71 extending in seriesbetween output terminals O1'. Switch S7 connects the junction of theseresistors to the lower end of emitter resistor R56 of the first stage toapply thereto a direct-current degenerative feedback signal.

Another advantage of the circuit of FIG. 4a resides in the reverse-biasof the base-emitter junction of the "off" output transistor during itsnon-conductive half of the cycle. This raises the breakdown voltage ofthe output transistor and also improves its high-frequency performanceby rapidly sweeping out the majority carriers from the base to permitthe transistor to be quickly turned off near the crossover point.

Referring now to FIG. 5a, there is shown a well-known prior circuitutilizing an output coupling capacitor and referred to above in thedescription of the prior art. The basic configuration of this circuitwas first disclosed by H. C. Lin, "Quasi-Complementary TransistorAmplifier," Electronics, Sept. 1956, pp. 173-175. The hot input terminalI11 is coupled by capacitor C26 to the base of transistor Q29 operatingClass A in the common-emitter mode with its emitter grounded as shown. Abias resistor R72 extends from the base to ground and the other inputterminal I12 is also grounded. Extending in series to the collector oftransistor Q29 from the B+ terminal of the power supply indicatedsymbolically by the battery symbol designated PS are collector loadresistors R73, R74, temperature-compensating bias diodes D16, D17 and abias resistor R75.

The drive stage is the quasi-complementary type and comprises an NPNtransistor Q30 and an PNP transistor Q32. The collector of transistorQ30 is connected to the B+ supply terminal and its base isdirect-coupled to the lower end of load resistor R74. A bias resistorR76 extends from the emitter of transistor Q30 to the output bus O.Connected to the latter is the emitter of transistor Q32 having its basedirect-coupled to the collector of transistor Q29. A collector loadresistor R77 extends from the collector of transistor Q32 to ground.

The output stage comprises a pair of NPN transistors Q32, Q33. The baseof transistors Q32 is direct-coupled to the emitter of drive transistorQ30, and the base of transistor Q33 is direct-coupled to the collectorof drive transistor Q32. The collector of output transistor Q31 isconnected to the B+ supply terminal and its emitter is connected tooutput bus O to which is also connected the collector of outputtransistor Q33 having its emitter grounded as shown.

Extending from output bus O to the junction of load resistors R73, R74of the first stage is the usual bootstrapping capacitor C27. A variableresistor in the form of a potentiometer P extends from output bus O tothe base of transistor Q29 to provide bias and direct-current feedbackto this stage.

Output bus O at the midpoint of the output stage Q31, Q33 is connectedby output coupling capacitor C28 to the hot output terminal O3. Theother output terminal O4 is grounded and the loudspeaker S is connectedbetween output terminals O3, O4. An alternating-current feedback networkcomprising a resistor R78 and phase-shift capacitor C29 extends fromoutput terminal O3 to the base of transistor Q29.

Some of the disadvantages of this prior art circuit have been mentionedbriefly above in the description of the prior art reference thereto willnow be made in conjunction with the specific structure disclosed in FIG.5a. Since transistor Q29 is the only stage which provides voltage gain,the amount of alternating-current negative feedback which may be appliedis limited in order not to reduce the sensitivity of the amplifier tothe point where an excessively large input signal is required for fullpower output. In order to provide more voltage gain and feedback, atleast one more common-emitter Class A stage is often added beforetransistor Q29 and within the forward transmission path of the outerfeedback loop.

The addition of a single common-emitter stage before transistor Q29 doesnot result in a substantial reduction in distortion because only amoderate increase in voltage gain is achieved and this is partiallyoffset by the distortion in the added stage. The gain increase ismoderate because the inner feedback network P to the base of transistorQ29 results in an extremely low input impedance seen by the added stagewhich is thus provided with a steep load-line. This loading of the addedstage also results in the generation of nonlinear distortion therein.

Therefore, two mutually direct-coupled stages are frequently addedbefore capacitor C26 and transistor Q29. Although sufficient gain topermit a substantial amount of feedback in the outer loop is therebymade available, other disadvantages arise from this expedient. Thereresult two additional stages of high-frequency roll-off and theaccompanying phase shifts reduce the stability margin of the outerfeedback loop. Also, the outer feedback network R78, C29 must now bereturned to the base of the initial added transistor resulting in a lowimpedance at the input terminals of the amplifier and thereby loadingdown the preamplifier, tuner or other preceding component.

In an effort to eliminate these disadvantages another attempted solutionis to direct-couple two initial common-emitter stages andcapacitor-couple the second of these stages to the drive stage. Thiseliminates the inner feedback network and the gain loss and distortionproduced thereby in the stage preceding the network.

These expedients, although providing sufficient negative feedback,nevertheless render the stability margins inadequate at both frequencyextremes. With respect to low frequencies, the forward transmission paththen includes two phase-shift producing roll-offs in the form ofcoupling capacitors C26 and C28. Separation of the two roll-offfrequencies is limited by the low impedance in the stages before andafter capacitor C26. With the application of sufficient feedback thepoles of the overall transfer function become complex so as to produce aspurious oscillatory response known as "breathing" when subjected totransient signals. With respect to high frequencies, coupling capacitorsC26 and C28 provide both series inductance and stray shunt capacitanceso as to substantially reduce the stability margin.

Furthermore, the interstage coupling capacitor changes its state ofcharge in response to overload signal peaks or saturation of thefollowing stage. During the finite time for recovery by capacitor C26 ofits original state of charge the bias on transistor Q29 is disturbed soas to cause distortion or even complete "blocking" of the amplifier.

Other disadvantages of output coupling capacitor C28 are the generationof transient distortion due to a disturbance in the bias condition of anearly stage and produced by a low-frequency instability or by a heavypulse having a substantial direct-current component which changes thecharge condition of capacitor C28. The latter further provides areactive load resulting in an elliptical load-line so that in responseto low-frequency transient signals the operating point of outputtransistor Q31 or Q33 may enter a region of simultaneously high voltageand current so as to result in "second breakdown" of the transistor.

Several terms and phrases which appear throughout the claims are herebydefined as follows. The expressions "source of reference potential" and"reference node" refer to either ground or a node maintained at aregulated potential different from ground potential. The terms "network"and "circuit means" and similar expressions are generic to both activenetworks.Iadd., .Iaddend.including transistors and passive networksconsisting solely of passive components such as conductors, resistors,capacitors and inductors. The expression "direct-current-coupled" isgeneric to include not merely a direct coupling but any coupling whichhas a conductive or direct-current transmission path, and the couplingmay be either by way of a passive network or an active network includinga base-emitter junction or one or more transistor stages. The term"transistor" includes any semiconductor device capable of voltage and/orcurrent amplification, whether the device be a discrete component orpart of an integrated monolithic or hybrid circuit. The expression"single-ended push-pull stage" refers to either a complementary-symmetrystage wherein the transistors are of opposite polarity types or a stagewherein both transistors are of the same polarity type. The phrase"common-emitter stage" is intended to include either a single transistoror an emitter-coupled differential pair of transistors.

It is to be understood that the various forms of invention shown in thedrawings and described in detail above are merely illustrative and thatnumerous modifications thereof will readily occur to those skilled2,896,029, the art without departing from the scope of the invention asdelineated in the appended claims which are to be construed as broadlyas permitted by the prior art.

REFERENCES CITED IN PRIOR APPLICATION SER. NO. 388,399

United States Patents: 2,798,164, Stanley, Apr. 1957; 2,847,519,Aronson, Aug. 1958; 2,851,542, Lohman, Sept. 1958; 2,860,195, Stanley,Nov. 1958; 2,863,008, Keonjian, Dec. 1958; 2,896,029, Lin, July 1959;2,955,258, Wheatley, Oct. 1960; 3,018,445, Stone, Jan. 1962; 3,023,368,Erath, Feb. 1962; 3,042,875, Higginbotham, July 1962; 3,246,251,Sheppard, Apr. 1966.

I claim: .[.1. A transistor power amplifier comprising a single-endedpush-pull output stage including at least two transistors connected inseries at a midpoint of the stage, a split power connected to saidoutput stage and having a center-tap, a pair of output terminals, meansD.C.-coupling one of said output terminals to said center-tap, meansD.C.-coupling the other output terminal to said output stage midpoint, acomplementary-symmetry push-pull drive stage, means D.C.-coupling saiddrive stage to said output stage, amplification means, meansD.C.-coupling said amplification means to said drive stage, and a D.C.feedback network extending from said output stage to said amplicationmeans for maintaining said output terminals at substantially the sameD.C. potential..]. .[.2. An amplifier as recited in claim 1 wherein oneend of said feedback network is D.C.-coupled to said other outputterminal, said amplification means having a negative feedback signalinjection node, the other end of said feedback network beingD.C.-coupled to said signal injection node so as to form a D.C. negativefeedback loop comprising said amplification means, drive stage, outputstage and feedback network..]. .[.3. An amplifier as recited in claim 2wherein said amplification means comprises a common-emitter stageincluding a transistor having an input circuit electrode and acollector, means D.C.-coupling said collector to said drive stage, saidamplifier having a ground, resistance means having one end connected tosaid electrode, and zener diode means maintaining the other end of saidresistance means at a substantially fixed potential with respect to saidground..]. .[.4. An amplifier as recited in claim 2 wherein saidamplification means comprises a transistor having a base constitutingsaid feedback signal injection node, said other end of said feedbacknetwork being D.C.-coupled to said base..]. .[.5. In combination, a pairof amplifiers each as recited in claim 2, and means for connecting saidamplifiers alternatively in either a stereo mode, or a parallel mode ora series mode, in said stereo mode each of said amplifiers havingrespective independent input and output terminals so as to constitutetwo independent channels, in said parallel mode said amplifiers having acommon input and having their outputs mutually D.C.-coupled to eachother and to the load, in said series mode one of said amplifiers havingphase reversal means so as to operate in opposite phase to the otheramplifier and said amplifiers each having its respective outputD.C.-coupled to a respective opposite end of the load..]. .[.6. Atransistor power amplifier as recited in claim 1 wherein saidamplification means comprises at least one transistor operating in thecommo-emitter mode and having a base and an emitter, said drive stagecomprising at least two complementary transistors each having anemitter, an A.C. ground, means connecting said amplification transistoremitter to said A.C. ground, means connecting said drive transistoremitters to said output stage midpoint, said D.C. feedback networkincluding a passive impedance having one end D.C.-coupled to said outputstage and its other end D.C.-coupled to said amplification transistorbase..].
 7. .[.In combination, an amplifier as recited in claim 1,.]..Iadd.A transistor power amplifier comprising a single-ended push-pulloutput stage including at least two transistors connected in series at amidpoint of the stage, a split power supply connected to said outputstage and having a center-tap, a pair of output terminals, meansD.C.-coupling one of said output terminals to said center-tap, meansD.C.-coupling the other output terminal to said output stage midpoint, acomplementary-symmetry push-pull drive stage, means D.C.-coupling saiddrive stage to said output stage, amplification means, meansD.C.-coupling said amplification means to said drive stage, and a D.C.feedback network extending from said output stage to said amplificationmeans for maintaining said output terminals at substantially the sameD.C. potential, .Iaddend.preamplifier means connected in cascade withsaid amplifier, an inner negative feedback loop including said amplifierfor raising the high-frequency cutoff of said amplifier to apredetermined frequency, said preamplifier means comprising a transistorstage preceding said inner feedback loop, said transistor stage having ahigh-frequency cutoff substantially lower than said predeterminedfrequency, and an outer negative feedback loop including said amplifierand said preamplifier transistor stage. .[.8. A transistor poweramplifier as recited in claim 1 wherein said amplification meanscomprises at least a first transistor of one polarity type and a secondtransistor of complementary type and each transistor having a collectorand a base, a network D.C.-coupling the first transistor collector tothe second transistor base, a ground, a bias reference node maintainedat a potential relatively fixed with respect to said ground andindependent of potential variations in said power supply, bias meansconnecting said bias reference node to said first transistor base tosupply bias current to the latter, said drive stage comprising at leasttwo complementary transistors each having an emitter and a base, networkmeans constituting D.C. transmission paths from said second transistorcollector to said drive transistor bases, and means connecting saiddrive transistor emitters to said output stage midpoint..]. .[.9. Incombination, an amplifier as recited in claim 1, preamplifier meansconnected in cascade with said amplifier, an inner negative feedbackloop including said amplifier for lowering the high-frequency cutoff ofsaid amplifier to a predetermined frequency, said preamplifier meanscomprising a transistor stage preceding said inner feedback loop, saidtransistor stage having a high-frequency cutoff substantially higherthan said predetermined frequency, and an outer negative feedback loopincluding said amplifier and said preamplifier transistor stage..]..[.10. In combination, a pair of amplifiers each as recited in claim 1,and means for connecting said amplifiers alternatively in either aparallel mode or a series mode, in said parallel mode said amplifiershaving a common input and having their outputs mutually D.C.-coupled toeach other and to the load, in said series mode one of said amplifiershaving phase reversal means so as to operate in opposite phase to theother amplifier and said amplifiers each having its respective outputD.C.-coupled to a respective opposite end of the load..]. .[.11. Atransistor power amplifier as recited in claim 1 wherein saidamplification means comprises a first transistor of predeterminedpolarity type and a second transistor of opposite polarity type, saidfirst and second transistors each having a collector, a base and anemitter, bias means maintaining said first transistor base at arelatively fixed direct-current quiescent potential independent ofpotential variations of said power supply, an auxiliary power supplyhaving a supply terminal, a load impedance extending from said auxiliarysupply terminal to said first transistor collector, a networkdirect-current coupling said first transistor collector to said secondtransistor base, means direct-current coupling said second transistorcollector to said drive stage, an alternating-current ground, meansconnecting said second transistor emitter to said alternating-currentground, and means transmitting the D.C. feedback network signal to saidfirst transistor emitter..]. .[.12. A transistor power amplifier asrecited in claim 11 wherein said output stage comprises at least twooutput transistors each having a base, an emitter and a collector, meansD.C.-coupling the emitter of one output transistor and the collector ofthe other output transistor to said output terminal, said complementarydrive stage comprising at least two drive transistors of oppositepolarity type and each having a base, an emitter and a collector,circuit means providing a D.C. transmission path from the emitter of oneof said drive transistors to the base of one of said output transistors,circuit means providing a D.C. transmission path from the collector ofthe other drive transistor to the base of the other output transistor,and circuit means providing D.C. transmission path from said collectorof said second transistor to the respective bases of said drivetransistors..]. .[.13. A transistor power amplifier as recited in claim1 for driving a loudspeaker and reproduction thereby without audibledistortion by said amplifier of a high-fidelity music signal fed theretoand wherein said drive stage comprises a pair of transistors each havinga base and an emitter, said D.C.-coupling means including network meanshaving D.C. transmission paths from said amplification means to saiddrive stage bases, and means conductively connecting said drive stageemitters to said other output terminal whereby in response to ashort-circuit failure of one of said output transistors one of saiddrive stage transistors will conduct current from the power supply in apath bypassing the loudspeaker so as to prevent damage to the latter..]..[.14. A transistor power amplifier as recited in claim 1 wherein saidoutput stage comprises a pair of complementary transistors, said drivestage comprising a pair of complementary transistors each direct-currentcoupled to a respective one of said output transistors, secondamplification means connected in cascade between said first-recitedamplification means and said drive stage, and an inner second feedbacknetwork extending around said output and drive stages and said secondamplification means..]. .[.15. A transistor power amplifier as recitedin claim 14 and comprising means biasing each of said output stagetransistors in the cutoff region for Class B operation thereof, saidinner feedback network reducing the crossover distortion generated bythe Class B output stage..]. .[.16. A transistor power amplifier asrecited in claim 15 wherein said inner feedback network providessubstantially unity gain in the subcircuit consisting of the secondamplification means and drive and output stages..]. .[.17. A transistorpower amplifier as recited in claim 16 wherein said second amplificationmeans comprises a transistor having a base, a collector and an emitter,first network means coupling said first-recited amplification means tosaid base, and second network means coupling said collector to saiddrive stage, said inner feedback network being connected to saidemitter, whereby said subcircuit effectively operates in theemitter-follower mode with the potential of said other output terminalsubstantially following that of said second amplification meanstransistor emitter..]. .[.18. A transistor power amplifier as recited inclaim 17 wherein said second amplification means transistor is of apredetermined polarity type, said second amplification means including asecond transistor of a polarity type opposite thereto and having a basecoupled to the collector thereof and a collector coupled to said drivestage..]. .[.19. A transistor power amplifier as recited in claim 18wherein each of said output stage transistors comprises a base and anemitter connected to said output stage midpoint, each of said drivestage transistors comprising a base and an emitter connected to arespective one of said output stage transistor bases, said collector ofsaid second transistor of said second amplification means beingconnected to said drive stage transistor bases..].
 20. A transistorpower amplifier .[.as recited in claim 1.]. for reproduction withoutaudible distortion by said amplifier of a high-fidelity music signal and.Iadd.comprising a single-ended push-pull output stage including atleast two transistors connected in series at a midpoint of the stage, asplit power supply connected to said output stage and having acenter-tap, a pair of output terminals, means D.C.-coupling one of saidoutput terminals to said center-tap, means D.C.-coupling the otheroutput terminal to said output stage midpoint, a complementary-symmetrypush-pull drive stage, means D.C.-coupling said drive stage to saidoutput stage, amplification means, means D.C.-coupling saidamplification means to said drive stage, and a D.C. feedback networkextending from said output stage to said amplification means formaintaining said output terminals at substantially the same D.C.potential, and .Iaddend.wherein said amplification means comprises atleast a first transistor of one polarity type and a second transistor ofcomplementary type and each transistor having a collector, a base and anemitter, network means constituting a D.C. signal transmission path fromthe first transistor collector to the second transistor base, a ground,bias means maintaining said first transistor base at a quiescent D.C.potential relatively fixed with respect to said ground, meansA.C.-coupling said second transistor emitter to said ground, saidfeedback network transmitting a feedback signal to vary the potential atsaid first transistor emitter.Iadd., said feedback network beingconnected to said first transistor emitter, and means for transmittingan input signal to said first transistor base.Iaddend.. .[.21. Atransistor power amplifier as recited in claim 20 wherein said powersupply includes a pair of supply terminals of opposite polarities withrespect to ground, means for conducting current and a ripple componenttherewith to said supply terminals at varying voltage levels dependentupon the load regulation characteristics of said power supply asincreased load current is drawn therefrom, and said bias meansmaintaining the quiescent potential of said base at a substantiallyfixed voltage with respect to said ground and independent of said ripplecomponent and load regulation characteristic of the power supply..]..[.22. A transistor power amplifier as recited in claim 1 for driving aloudspeaker and reproduction thereby without audible distortion by saidamplifier of a high-fidelity music signal fed thereto and wherein saidamplification means comprises at least a first transistor of onepolarity type and a second transistor of complementary type and eachtransistor having a collector, a base and an emitter, a networkD.C.-coupling the first transistor collector to the second transistorbase, an A.C. ground, a bias reference node maintained at a potentialrelatively fixed with respect to said ground and independent ofpotential variations in said power supply, bias means connecting saidbias reference node to said first transistor base to supply bias currentto the latter, means connecting said second transistor emitter to saidground, said feedback network transmitting a feedback signal to vary thepotential at said first transistor emitter, said drive stage comprisingat least two complementary transistors each having an emitter and abase, network means constituting D.C. transmission paths from saidsecond transistor collector to said drive transistor bases, and meansconductively connecting said drive transistor emitters to said outputstage midpoint whereby in response to a short-circuit failure of one ofsaid output transistors one of said drive transistors will conductcurrent from the power supply in a path bypassing the loudspeaker so asto prevent damage to the latter..]. .[.23. A transistor power amplifieras recited in claim 22 wherein said amplification means comprises atleast a third transistor having a base and an emitter, means connectingsaid third transistor emitter to said first transistor emitter, commonimpedance means connecting said connected first and third transistoremitters to A.C. ground whereby said first and third transistors coactin the differential amplifier mode, said feedback network having apassive component including an impedance extending from said outputstage to said third transistor base and said feedback network furtherhaving an active component including said third transistor to transmitsaid feedback signal to said first transistor emitter, a D.C. ground,means D.C.-coupling said power supply center-tap to said D.C. ground,said bias reference node being at D.C. ground potential, said bias meanscomprising resistor means extending from said reference node to saidfirst transistor base, a second single-ended push-pull output stage, asecond complementary-symmetry push-pull drive stage D.C.-coupled betweensaid amplification means and said second output stage, and switchingmeans for disconnecting said one output terminal from said center-tapand connecting said one output terminal to said second output stagewhereby said output stages may operate in series..]. .[.24. A transistorpower amplifier as recited in claim 22 and having a D.C. ground, saidbias reference node being at the potential of said D.C. ground, saidbias means comprising a relative network extending from said biasreference node to said first transistor base, a feedback signal pickoffnode, and a voltage divider network connected between said power supplyand said output stage midpoint to impart to said feedback pickoff nodepotential variations coextensive with those of said output stagemidpoint, said feedback network transmitting said feedback signal fromsaid pickoff node to said first transistor emitter..]. .[.25. Atransistor power amplifier as recited in claim 1 wherein saidamplification means comprises a transistor operating in thecommon-emitter mode and having a collector, base and emitter, meansD.C.-coupling said collector to said drive stage, said feedback networkbeing connected to said emitter, a ground, a semiconductor device havinga diode junction with two electrodes connected thereto, means connectingone of said electrodes to said ground, and means connecting the otherelectrode to said base to maintain the latter at a substantially fixedreference potential..]. .[.26. A transistor power amplifier as recitedin claim 25 wherein said semiconductor device is a zener diode..]..[.27. A transistor power amplifier as recited in claim 25 wherein saidsemiconductor device is a transistor operating in the emitter-followermode and having a base and emitter constituting said electrodes,resistive means connecting said emitter-follower transistor base to saidground, input signal means connected to said emitter-follower transistorbase, and means D.C.-coupling said emitter-follower transistor emitterto said common-emitter transistor base to provide a D.C. bias theretoand to transmit said signal thereto..]. .[.28. A transistor poweramplifier as recited in claim 27 wherein said amplification meanscomprises a second common-emitter transistor of a polarity type oppositethat of said first-recited common-emitter transistor and having a baseand a collector, means D.C.-coupling said first-recited common-emittertransistor collector to said second common-emitter transistor base, andmeans D.C.-coupling said second common-emitter transistor collector tosaid drive stage..]. .[.29. A transistor power amplifier as recited inclaim 1 wherein said amplification means comprises a transistor having acollector, said drive stage including a transistor having a base,network means forming a direct-current signal transmission path fromsaid collector to said base, said power supply having a terminal at apredetermined polarity and potential, said output stage being connectedto said power supply terminal, an auxiliary power supply having aterminal at a potential of the same polarity as and of a greatermagnitude than said first-recited supply terminal and substantiallyisolated from any ripple present at the latter, and a collector loadimpedance network extending from said collector to said auxiliary powersupply terminal..]. .[.30. A transistor power amplifier comprising afirst stage including a transistor having a base electrode, an emitterelectrode and a collector, means transmitting an input signal to saidbase electrode, a ground, a potential source maintained at asubstantially fixed predetermined voltage with respect to said groundindependent of variations in power supplied to or drawn from saidamplifier, means connecting one of said electrodes to said potentialsource, an output stage, circuit means providing a D.C. signaltransmission path from said collector to said output stage, an outputterminal adapted to be connected to a loudspeaker or other load, apassive direct-current feedback network extending from said output stageto one of said electrodes, and means D.C.-coupling said output stage tosaid output terminal..]. .[.31. A transistor power amplifier as recitedin claim 30 wherein said circuit means includes a drive stage comprisingat least two complementary transistors each having an emitter and abase, network means constituting D.C. transmission paths from said firststage transistor collector to said drive transistor bases, and meansconnecting said drive transistor emitters to said output terminal..]..[.32. A transistor power amplifier as recited in claim 30 foramplifying without audible distortion a high-fidelity music signal andcomprising a power supply including a pair of supply terminals ofopposite polarities with respect to said ground, means for conductingcurrent and a ripple component therewith to said supply terminals atvarying voltage levels dependent upon the load regulation characteristicof said power supply as increased load current is drawn therefrom, andbiasing means including said potential source for maintaining thequiescent potential of said one electrode at a substantially fixedvoltage with respect to said ground and independent of said ripplecomponent and load regulation characteristic of the power supply, andmeans connecting said output stage to said power supply terminals..]..[.33. A transistor power amplifier as recited in claim 30 wherein saidpotential source is said ground, said connecting means comprising asecond transistor having a base and an emitter, means connecting saidlast-recited base to said ground, and means connecting said last-recitedemitter to said one electrode of said first-recited transistor..]..[.34. A transistor power amplifier as recited in claim 33 wherein saidinput signal is transmitted by said second transistor to said baseelectrode, said last-recited emitter being direct-current coupled tosaid base electrode, said passive feedback network being direct-currentcoupled to said emitter electrode..].
 35. A transistor power amplifier.[.as recited in claim 30 and.]. comprising .Iadd.a first stageincluding a transistor having a base electrode, an emitter electrode anda collector, means transmitting an input signal to said base electrode,a ground, a potential source maintained at a substantially fixedpredetermined voltage with respect to said ground independent ofvariations in power supplied to or drawn from said amplifier, meansconnecting one of said electrodes to said potential source, an outputstage, circuit means providing a D.C. signal transmission path from saidcollector to said output stage, an output terminal adapted to beconnected to a loudspeaker or other load, a passive direct-currentfeedback network extending from said output stage to one of saidelectrodes, and means D.C.-coupling said output stage to said outputterminal, .Iaddend.a second stage including a transistor of a polaritytype complementary to that of said first stage having a collector, abase and an emitter, a network D.C.-coupling the first stage transistorcollector to the second stage transistor base, said potential sourcebeing connected to said first stage transistor base to supply biascurrent to the latter, means A.C.-coupling said second stage transistoremitter to said ground, said feedback network transmitting a feedbacksignal to vary the potential at said first transistor emitter, a drivestage comprising at least two complementary transistors each having anemitter and a base, network means constituting D.C. transmission pathsfrom said second stage transistor collector to said drive stagetransistor bases, and means connecting said drive stage transistoremitters to said output terminal.
 36. A transistor power amplifier asrecited in claim 35 and comprising a third transistor having acollector, a base and an emitter, means connecting said third transistoremitter to said first stage transistor emitter, common impedance meanshaving one end connected to said connected emitters, means A.C.-couplingthe other end of said impedance means to said ground whereby said firststage transistor and said third transistor coact in the differentialamplifier mode, said feedback network having a passive componentincluding an impedance extending from said output stage to said thirdtransistor base and said feedback network further having an activecomponent including said third transistor to transmit said feedbacksignal to said first stage transistor emitter, means D.C.-coupling saidpower supply center-tap to said ground, said bias reference node beingat D.C. ground potential, a second output stage, and network means forselectably connecting said second output stage either in parallel or inseries with said first output stage. .[.37. A transistor power amplifieras recited in claim 35, said potential source being at the potential ofsaid ground, a resistive network extending from potential source to saidfirst transistor base, a feedback signal pickoff node, and a voltagedivider network connected to said output stage midpoint to impart tosaid feedback pickoff node potential variations coextensive with thoseof said output stage midpoint, said feedback network transmitting saidfeedback signal from said pickoff node to said first stage transistoremitter..]. .[.38. A power amplifier comprising a first amplifier stageincluding an input circuit and an output circuit, an output stage havinga first output terminal for direct-current coupling to one terminal of aloudspeaker or other load, a power supply connected to said outputstage, a second output terminal for direct-current coupling to the otherterminal of said load, first circuit means direct-current coupling saidfirst amplifier stage output circuit to said output stage to drive thelatter, a substantially fixed source of reference potential of amagnitude substantially independent of variations in the potential ofsaid power supply, second circuit means connected to said referencepotential source and said first amplifier stage input circuit to supplybias current thereto at a relatively fixed quiescent potential so as toadjust the quiescent potential of said first output terminal to that ofsaid second output terminal, a node having a direct-current potentialvarying substantially proportionately to that of said first outputterminal, and a passive feedback impedance direct-current coupledbetween said node and said first amplifier stage, thereby providing adirect-current negative feedback loop to reduce any tendency of thequiescent potential of said first output terminal to vary from that ofsaid second output terminal..]. .[.39. A power amplifier .[.as recitedin claim 38.]. .Iadd.comprising a first amplifier stage including aninput circuit and an output circuit, an output stage having a firstoutput terminal for direct-current coupling to one terminal of aloudspeaker or other load, a power supply connected to said outputstage, a second output terminal for direct-current coupling to the otherterminal of said load, first circuit means direct-current coupling saidfirst amplifier stage output circuit to said output stage to drive thelatter, a substantially fixed source of reference potential of amagnitude substantially independent of variations in the potential ofsaid power supply, second circuit means connected to said referencepotential source and said first amplifier stage input circuit to supplybias current thereto at a relatively fixed quiescent potential so as toadjust the quiescent potential of said first output terminal to that ofsaid second output terminal, a node having a direct-current potentialvarying substantially proportionately to that of said first outputterminal, and a passive feedback impedance direct-current coupledbetween said node and said first amplifier stage, thereby providing adirect-current negative feedback loop to reduce any tendency of thequiescent potential of said first output terminal to vary from that ofsaid second output terminal, and .Iaddend.wherein said first amplifierstage comprises a first transistor of one polarity type and having abase and an emitter in said input circuit and a collector in said outputcircuit, said first circuit means including a second transistor ofopposite polarity type and having a base and a collector, a firstnetwork direct-current coupling said first transistor collector to saidsecond transistor base, a second network direct-current coupling saidsecond transistor collector to said output stage, said second circuitmeans being connected to said first transistor base to maintain thelatter at a substantially fixed quiescent potential, and said feedbackimpedance being connected to said first transistor emitter. .[.40. Apower amplifier as recited in claim 38 wherein said source of referencepotential comprises a zener diode..]. .[.41. A power amplifier asrecited in claim 38 wherein said source of reference potential comprisesan emitter-follower preamplifier stage including a transistor having abase and an emitter, a reference node maintained at a relatively fixedpotential, means direct-current coupling said node to said base, andmeans direct-current coupling said emitter to said first amplifier stageinput circuit..]. .[.42. A transistor power amplifier comprising a splitpower supply having a center-tap and two supply terminals of oppositepolarity, a single-ended push-pull output stage including a pair oftransistors connected in series at a junction and each having oneelectrode connected to a respective one of said supply terminals andanother electrode, a first output terminal for connection to one end ofa speaker or other load, first circuit means connecting said otherelectrodes of said output transistors to said first output terminal, asecond output terminal for connection to the other end of said load,means coupling said second output terminal to said supply center-tap, avoltage-amplification stage including a transistor having adirect-current feedback signal injection node and a collector, adirect-current feedback network for injecting into said node a feedbacksignal proportional to the direct-current potential of said junction,second circuit means direct-current coupling said voltage-amplificationtransistor collector to said output stage, said voltage-amplificationtransistor having a base electrode and an emitter electrode, a biassupply node having a relatively fixed voltage, a bias impedanceextending from said bias supply node to one of saidvoltage-amplification transistor electrodes to supply bias current tosaid one electrode and to maintain the latter at an approximatelyconstant direct-current potential, said other voltage-amplificationtransistor electrode constituting said feedback injection node andhaving said feedback network direct-current coupled thereto..]. .[.43. Atransistor power amplifier as recited in claim 42 and comprising a zenerdiode having an electrode, said bias supply node being said zener diodeelectrode..]. .[.44. A transistor power amplifier as recited in claim 42wherein said second circuit means comprises a secondvoltage-amplification transistor direct-current coupled to saidfirst-recited voltage-amplification transistor, and a push-pullcomplementary-symmetry drive stage direct-current coupled to said secondvoltage-amplification transistor and said output stage..].
 45. Atransistor power amplifier .[.as recited in claim 44.]. .Iadd.comprisinga split power supply having a center-tap and two supply terminals ofopposite polarity, a single-ended push-pull output stage including apair of transistors connected in series at a junction and each havingone electrode connected to a respective one of said supply terminals andanother electrode, a first output terminal for connection to one end ofa speaker or other load, first circuit means connecting said otherelectrodes of said output transitors to said first output terminal, asecond output terminal for connection to the other end of said load,means coupling said second output terminal to said supply center-tap, avoltage-amplification stage including a transistor having adirect-current feedback signal injection node and a collector, adirect-current feedback network for injecting into said node a feedbacksignal proportional to the direct-current potential of said junction,second circuit means direct-current coupling said voltage-amplificationtransistor collector to said output stage, said voltage-amplificationtransistor having a base electrode and an emitter electrode, a biassupply node having a relatively fixed voltage, a bias impedanceextending from said bias supply node to one of saidvoltage-amplification transistor electrodes to supply bias current tosaid one electrode and to maintain the latter at an approximatelyconstant direct-current potential, said other voltage-amplificationtransistor electrode constituting said feedback injection node andhaving said feedback network direct-current coupled thereto, and whereinsaid second circuit means comprises a second voltage-amplificationtransistor direct-current coupled to said first-recitedvoltage-amplification transistor, and a push-pull complementary-symmetrydrive stage direct-current coupled to said second voltage-amplificationtransistor and said output stage, and .Iaddend.wherein saidvoltage-amplification transistors are respectively of opposite polaritytypes with said second transistor having a base direct-current coupledto said collector of said first transistor, said bias impedanceextending to said base electrode, said emitter electrode constitutingsaid feedback injection node. .[.46. A transistor power amplifier asrecited in claim 45 and comprising an auxiliary power supply having anoutput substantially independent of ripple n said first-recited powersupply, said bias supply node and said voltage-amplification transistorsbeing connected to said auxiliary power supply..]. .[.47. A transistorpower amplifier comprising a single-ended push-pull output stage havinga first output terminal for direct-current coupling said stage to oneend of a loudspeaker or other load, a power supply connected to saidoutput stage, a second output terminal for direct-current coupling theother end of said load to said power supply, a preceding amplifyingstage comprising a transistor having a base electrode and an emitterelectrode, an active network direct-current coupling said precedingstage to said output stage, a bias network maintaining one of saidelectrodes at a relatively fixed direct-current quiescent potentialindependent of variations in the load current or in the power supplypotential, and a negative feedback network responsive to the potentialof said first output terminal and direct-current coupled to the otherelectrode for applying thereto a potential proportional to thedirect-current quiescent potential of said output terminal to maintainsaid quiescent potential substantially equal to that of said secondoutput terminal..]. .[.48. A transistor power amplifier as recited inclaim 47 for reproduction without audible distortion by said amplifierof a high-fidelity music signal and wherein said transistor is of apredetermined polarity type, said active network comprising a secondtransistor of the opposite polarity type and having a base, saidfirst-recited transistor having a collector, first circuit meansdirect-current coupling said last-recited collector to said base of thesecond transistor, said active network including second circuit meansdirect-current coupling said second transitor to said output stage..]..[.49. A transistor power amplifier as recited in claim 48 wherein saidbias network maintains the base electrode of said first-recitedtransistor of said fixed direct-current potential, said feedback networkbeing direct-current coupled to the emitter electrode of saidfirst-recited transistor..].
 50. A transistor power amplifier .[.asrecited in claim 49.]. .Iadd.for reproduction without audible distortionby said amplifier of a high-fidelity music signal and comprising asingle-ended push-pull output stage having a first output terminal fordirect-current coupling said stage to one end of a loudspeaker or otherload, a power supply connected to said output stage, a second outputterminal for direct-current coupling the other end of said load to saidpower supply, a preceding amplifying stage comprising a transistorhaving a base electrode and an emitter electrode, an active networkdirect-current coupling said preceding stage to said output stage, abias network maintaining one of said electrodes at a relatively fixeddirect-current quiescent potential independent of variations in the loadcurrent or in the power supply potential, and a negative feedbacknetwork responsive to the potential of said first output terminal anddirect-current coupled to the other electrode for applying thereto apotential proportional to the direct-current quiescent potential of saidfirst output terminal to maintain said quiescent potential substantiallyequal to that of said second output terminal, and wherein saidtransistor is of a predetermined polarity type, said active networkcomprising a second transistor of the opposite polarity type and havinga base, said first-recited transistor having a collector, first circuitmeans direct-current coupling said last-recited collector to said baseof the second transistor, said active network including second circuitmeans direct-current coupling said second transistor to said outputstage, and wherein said bias network maintains the base electrode ofsaid first-recited transistor at said fixed direct-current potential,said feedback network being direct-current coupled to the emitterelectrode of said first recited transistor, and .Iaddend.wherein saidsecond circuit means comprises a push-pull complementary-symmetry drivestage, third circuit means direct-current coupling said drive stage tosaid output stage, and fourth circuit means direct-current coupling saidsecond transistor to said drive stage. .[.51. A transistor poweramplifier as recited in claim 50 wherein said bias network comprises azener diode having a substantially predetermined breakdown voltage, andresistive means extending from said zener diode to said baseelectrode..]. .Iadd.
 52. An amplifier as recited in claim 39 whereinsaid first amplifier stage comprises another transistor of the samepolarity type as said first transistor and having a base-emitter diodeincluding an emitter coupled to the emitter of said first transistor andalso including a base connected to said feedback impedance to transmit afeedback signal in the back direction through said diode to said firsttransistor emitter. .Iaddend. .Iadd.
 53. In combination, a pair ofamplifiers each as set forth in claim 45, and means for connecting saidamplifiers alternatively in either a separate mode or a series mode, insaid separate mode each of said amplifiers having respective independentinput and output terminals whereby said pair of amplifiers constitutetwo independent channels, in said series mode said amplifiers eachhaving its respective output direct-current-coupled to a respectiveopposite end of the load, means for transmitting respective differentinput signals to said independent input terminals when said amplifiersare connected in the separate mode, and means for transmitting a firstinput signal to the input terminals of a first of said amplifiers andfor transmitting to the input terminals of the second amplifier a secondinput signal substantially identical to said first input signal butoppositely phased with respect thereto when said amplifiers areconnected in the series mode. .Iaddend. .Iadd.
 54. An amplifier asrecited in claim 47 and further comprisinga quasi-output stage incascade with and between said drive stage and said output stage, saidquasi-output stage including resistive means for conducting outputcurrent from said quasi-output stage to said output terminal and then tothe load at low amplifier signal levels, said resistive means having aresistance of an order of magnitude approximately the same as the orderof magnitude of the resistance of the load, means normally biasing saidoutput stage to the cutoff state at low amplifier signal levels, andmeans connecting said output stage to said resistive means for drivingsaid output stage to the active state so as to supply power to the loadat high amplifier signal levels. .Iaddend. .Iadd.
 55. An amplifier asrecited in claim 54 wherein said quasi-output stage comprises a pair oftransistors, said output stage comprising a pair of transistors each ofthe same polarity type as a respective one of said quasi-output stagetransistors, each of said quasi-output stage transistors having anemitter, each of said output stage transistors having a basedirect-coupled to a respective emitter of one of said quasi-output stagetransistors, said resistive means comprising a pair of resistors eachconnected to a respective emitter of one of said quasi-output stagetransistors. .Iaddend. .Iadd.
 56. An amplifier as set forth in claim 47and further comprising a second drive stage in cascade with and betweensaid first-recited drive stage and said output stage, said second drivestage comprising four transistors, said output stage comprising fourtansistors connected in series, each of said output stage transistorshaving a base, each of said second drive stage transistors beingdirect-coupled to the base of a respective one of said output stagetransistors, two of said second drive stage transistors having a base,and feedback means driven by the amplifier output for driving said twosecond drive stage transistor bases. .Iaddend. .Iadd.
 57. An amplifieras set forth in claim 56 wherein said feedback means comprises a pair ofvoltage divider networks each connected between the amplifier output anda respective one of the power supply terminals, each of said voltagedivider networks being direct-current-coupled to a respective one ofsaid two second drive stage transistor bases for driving the latter..Iaddend. .Iadd.
 58. An amplifier as recited in claim 45 wherein saidfirst-recited voltage-amplification stage comprises another transistorhaving an emitter coupled to the emitter of said first-recitedvoltage-amplification transistor and having a base-emitter diodejunction, said feedback network including said diode junction forinjecting said feedback signal into said emitter electrode constitutingsaid feedback signal injection node, and an input terminal connected tosaid base electrode of said first-recited voltage amplification stagetransistor. .Iaddend..Iadd.
 59. An amplifier as recited in claim 58wherein said another transistor has a base electrode, said feedbacknetwork comprising at least a first resistor, a second resistor and acapacitor, said first feedback network resistor having one end connectedto the amplifier output and its other end connected to said baseelectrode, a ground, said second feedback network resistor and saidcapacitor being connected in series between said base electrode and saidground. .Iaddend. .Iadd.
 60. An amplifier as recited in claim 59 whereinsaid bias supply node is at ground potential, said bias impedancecomprising at least a resistor extending from said base electrode toground. .Iaddend..Iadd.
 61. An amplifier as recited in claim 45 andcomprising an input terminal connected to said base electrode of saidfirst-recited voltage-amplification stage transistor, said first-recitedvoltage-amplification stage comprises another transistor having anemitter coupled to the emitter of said first-recitedvoltage-amplification transistor and having a base-emitter diode, saidfeedback network including said diode for transmitting said feedbacksignal in the feedback direction to said emitter electrode constitutingsaid feedback injection node, said drive stage comprising at least twodrive transistors each having an emitter connected to the amplifieroutput whereby the drive and output stages operate together in theemitter-follower mode so as to present an inherently high inputimpedance as seen looking into the drive stage, said secondvoltage-amplification transistor base presenting an inherently low inputimpedance to the preceding first-recited voltage-amplificationtransistor, whereby the frequency response of the secondvoltage-amplification transistor rolls off at a much lower frequencythan the other stages of the amplifier. .Iaddend. .Iadd.
 62. Anamplifier as recited in claim 45, said first voltage-amplification stagetransistor having a quiescent base-emitter potential difference betweenthe base and emitter electrodes thereof, a first resistive meansassociated with and extending between said base electrode and ground andhaving a first quiescent voltage thereacross, a second resistive meansassociated with said emitter electrode and constituting at least part ofsaid feedback network and having one end connected to said first outputterminal and having a second quiescent voltage thereacross, said secondoutput terminal being substantially at ground potential, network meansfor maintaining the quiescent potential of one of said electrodes at alevel displaced from ground potential by an amount substantially equalto the sum of said quiescent base-emitter potential difference and saidquiescent voltage across said resistive means associated with said otherelectrode to maintain the quiescent voltage of said first outputterminal at substantially the same potential as said second outputterminal, and means for direct-current coupling the other end of saidsecond resistive means to said emitter electrode to transmit saidfeedback signal to the latter in the back direction and the voltageacross said coupling means being of a magnitude to enable said firstoutput terminal quiescent voltage to be maintained at substantially thesame potential as said second output terminal. .Iaddend. .Iadd.
 63. Anamplifier as recited in claim 62 wherein said feedback network comprisesat least a first resistor, a second resistor and a capacitor, said firstresistor having one end coupled to said first output terminal, saidsecond resistor and said capacitor being connected in series betweensaid other end of said first resistor and ground, and wherein said drivestage presents a relatively high input impedance to said secondtransistor, and said second transistor presents a relatively low inputimpedance to said first transistor, whereby the high-frequency responseof said second transistor rolls off at a lower frequency than that ofsaid first transistor to increase the high-frequency stability margin ofthe amplifier, said feedback network including a feedback resistor fortransmitting a feedback signal in the back direction toward said emitterelectrode of said first voltage-amplification stage transistor, andconductive means distinct from said feedback resistor and connected tosaid emitter electrode of said first voltage-amplification stagetransistor for conducting at least the major portion of the emittercurrent thereof. .Iaddend. .Iadd.
 64. A transistor power amplifier asset forth in claim 50 whereinsaid output stage comprises a pair ofoutput transistors of the same polarity type, each of said outputtransistors having a base, an emitter and a collector, means connectingthe collector of a first of said output transistors to said powersupply, means connecting the emitter of said first output transistor tosaid output terminal, means connecting the collector of the secondoutput transistor to said output terminal, means connecting the emitterof the second output transistor to the power supply, meansdirect-current-coupling the emitter of a first of said drive stagetransistors to the base of said first output transistor, and meansdirect-current-coupling the collector of the second drive stagetransistor to the base of said second output transistor. .Iaddend..Iadd.
 65. In combination, a pair of amplifiers each as set forth inclaim 50, and means for connecting said amplifiers alternatively ineither a separate mode or a series mode, in said separate mode each ofsaid amplifiers having respective independent input and output terminalswhereby said pair of amplifiers constitute two independent channels, insaid series mode said amplifiers each having its respective outputdirect-current-coupled to a respective opposite end of the load, meansfor transmitting respective different input signals to said independentinput terminals when said amplifiers are connected in the separate mode,and means for transmitting a first input signal to the input terminalsof a first of said amplifiers and for transmitting to the inputterminals of the second amplifier a second input signal substantiallyidentical to said first input signal but oppositely phased with respectthereto when said amplifiers are connected in the series mode. .Iaddend..Iadd.
 66. An amplifier as recited in claim 50 wherein saiddirect-current-coupling renders the amplifier capable of transmitting alarge low-frequency transient signal, and a dissipation-limitingprotective circuit for preventing the respective operating points of theoutput transistors from entering a region of high dissipation. .Iaddend..Iadd.
 67. An amplifier as recited in claim 66 wherein said secondtransistor has an output network, said dissipation-limiting protectivecircuit including a pair of complementary transistors each having abase, an emitter and a collector, first means connecting said protectivecircuit transistor collectors to said output network of said secondcommon-emitter stage, resistive means for sensing the load current,second means connecting said bases and emitters of said protectivecircuit transistors to said resistive means, and a pair of resistorseach having one end connected to a respective one of said protectivecircuit transistor bases and its opposite end connected to a node atapproximately ground potential. .Iaddend. .Iadd.
 68. An amplifier asrecited in clam 67 wherein said first connecting means comprises a pairof diodes each having one end connected to said collector of arespective one of said protective circuit transistors and its oppositeend connected to said output network of the second transistor. .Iaddend..Iadd.
 69. An amplifier as recited in claim 50 and further comprising aquasi-output stage in cascade with and between said drive stage and saidoutput stage, said quasi-output stage including resistive means forconducting output current from said quasi-output stage to said outputterminal and then to the load at low amplifier signal levels, saidresistive means having a resistance of an order of magnitudeapproximately the same as the order of magnitude of the resistance ofthe load, means normally biasing said output stage to the cutoff stateat low amplifier signal levels, and means connecting said output stageto said resistive means for driving said output stage to the activestate so as to supply power to the load at high amplifier signal levels..Iaddend. .Iadd.
 70. An amplifier as recited in claim 69 wherein saidquasi-output stage comprises a pair of transistors, said output stagecomprising a pair of transistors each of the same polarity type as arespective one of said quasi-output stage transistors, each of saidquasi-output stage transistors having an emitter, each of said outputstage transistors having a base direct-coupled to a respective emitterof one of said quasi-output stage transistors, said resistive meanscomprising a pair of resistors each connected to a respective emitter ofone of said quasi-output stage transistors. .Iaddend. .Iadd.
 71. Anamplifier as set forth in claim 50 and further comprising a second drivestage in cascade with and between said first-recited drive stage andsaid output stage, said second drive stage comprising four transistors,said output stage comprising four transistors connected in series, eachof said output stage transistors having a base, each of said seconddrive stage transistors being direct-coupled to the base of a respectiveone of said output stage transistors, two of said second drive stagetransistors having a base, and feedback means driven by the amplifieroutput for driving said two second drive stage transistor bases..Iaddend. .Iadd.
 72. An amplifier as set forth in claim 71 wherein saidfeedback means comprises a pair of voltage divider networks eachconnected between the amplifier output and a respective one of the powersupply terminals, each of said voltage divider networks beingdirect-current-coupled to a respective one of said two second drivestage transistor bases for driving the latter. .Iaddend. .Iadd.
 73. Anamplifier as recited in claim 50 and comprising a diode connected tosaid emitter electrode for transmitting thereto in the feedbackdirection a feedback signal flowing through said feedback network, andan input terminal connected to said base electrode of said precedingamplifying stage transistor. .Iaddend. .Iadd.
 74. An amplifier asrecited in claim 73 and comprising another transistor having a base andan emitter constituting said diode, said last-recited emitter beingconnected to said first-recited emitter electrode, said feedback networkbeing direct-current coupled to said first-recited emitter electrodethrough said base and emitter of said another transistor. .Iaddend..Iadd.
 75. A transistor power amplifier for high-fidelity musicreproduction without audible distortion or listening fatigue underextended critical listening conditions and comprising a single-endedpush-pull output stage having a first output terminal for direct-currentcoupling said stage to one end of a loudspeaker or other load, a powersupply connected to said output stage, a second output terminal fordirect-current coupling the other end of said load to said power supply,a preceding amplifying stage comprising a transistor having a baseelectrode and an emitter electrode, an active network direct-currentcoupling said preceding stage to said output stage, a bias networkmaintaining one of said electrodes at a relatively fixed direct-currentquiescent potential independent of variations in the load current or inthe power supply potential, and a negative feedback network responsiveto the potential of said first output terminal and direct-currentcoupled to the other electrode for applying thereto a potentialproportional to the direct-current quiescent potential of said firstoutput terminal to maintain said quiescent potential substantially equalto that of said second output terminal, and wherein said precedingamplifying stage transistor is connected in the common-emitter mode,said active network includinga push-pull drive stage comprising a pairof complementary drive transistors each having an emitter, and meansconnecting said emitters to said output terminal whereby said drive andoutput stages together effectively operate in the emitter-follower mode,and wherein said output stage comprises a pair of output transistors ofthe same polarity type and each having a base and an emitter, one ofsaid drive transistors having an emitter direct-current coupled to thebase of a respective one of said output transistors, the other of saiddrive transistors having a collector direct-current coupled to the baseof the other output transistor, the emitter of said one outputtransistor being connected to said first output terminal, said powersupply having an ungrounded supply terminal, and conductive meansconnecting the emitter of said other output transistor to saidungrounded power supply terminal, and wherein said precedingcommon-emitter amplifying stage transistor has a collector and anemitter and is of a predetermined polarity type, a second common-emitteramplifying stage including a transistor of a polarity type opposite tothat of said preceding amplifying stage transistor and having a base anda collector, said first common-emitter transistor collector beingdirect-current-coupled to said second common-emitter transistor base,said second common-emitter transistor collector beingdirect-current-coupled to said drive stage, said bias networkmaintaining said base electrode of said first common-emitter precedingamplifying stage transistor at said relatively fixed direct-currentquiescent potential and including a resistor extending from said baseelectrode to ground, said negative feedback network beingdirect-current-coupled to said emitter electrode of the firstcommon-emitter preceding amplifying stage transistor, said feedbacknetwork including a feedback resistor for transmitting a feedback signalin the back direction toward said last-recited emitter electrode, andconductive means separate from said feedback resistor and connected tosaid emitter electrode for conducting at least a substantial portion ofthe emitter current of said first common-emitter preceding amplifyingstage transistor. .Iaddend. .Iadd.
 76. A transistor power amplifier forhigh fidelity music reproduction without audible distortion or listeningfatigue under extended critical listening conditions and comprisingasingle-ended push-pull output stage including at least two transistorsconnected in series at a midpoint of the stage, a split power supplyconnected to said output stage and having a center-tap, a pair of outputterminals, means D.C.-coupling one of said output terminals to saidcenter-tap, means D.C.-coupling the other output terminal to said outputstage midpoint, a complementary-symmetry push-pull drive stage, meansD.C.-coupling said drive stage to said output stage, amplificationmeans, means D.C.-coupling said amplification means to said drive stage,a D.C. feedback network extending from said output stage to saidamplification means for maintaining said output terminals atsubstantially the same D.C. potential, said amplification meanscomprising two common-emitter stages direct-current-coupled in cascade,a first of said common-emitter stages including at least one transistorof a predetermined polarity type and connected in the common-emitterphase-reversing mode and having a base, a collector and an emitter, thesecond of said common-emitter stages including a transistor of theopposite polarity type and connected in the common-emitterphase-reversing mode and having a base direct-current-coupled to saidfirst common-emitter stage transistor collector and having a collectordirect-current-coupled to said drive stage, said first common-emitterstage having a base bias network for maintaining said transistor basethereof at a relatively fixed direct-current quiescent potential, saidbase bias network comprising a resistor extending from said transistorbase to ground, said first common-emitter stage having a feedbackinjection node direct-current-coupled to said emitter to apply afeedback signal effectively in series with the input signal so as toraise the input impedance of said first common-emitter stage in responseto the injection of a negative feedback signal into said node, said D.C.feedback network being direct-current-coupled to said feedback injectionnode. .Iaddend. .Iadd.
 77. A direct-current-coupled amplifier as setforth in claim 76 and capable of transmitting a large low-frequencysignal, and further comprising a dissipation-limiting protective circuitto prevent the operating point of the output stage from entering atleast one of the regions of high dissipation. .Iaddend. .Iadd.
 78. Anamplifier as set forth in claim 77 wherein said dissipation-limitingprotective circuit comprisesa current-sensing resistor in series withthe load and for conducting and sensing the instantaneous load current,a protective transistor having a base-emitter junction and a collector,said drive stage comprising a pair of drive transistors, one of saiddrive transistors having a base, a subnetwork connected to said base fortransmitting an input signal to the latter whereby the A.C. signalvoltage at the base is substantially the same as that at a node of thesubnetwork, means connecting said protective transistor collector tosaid subnetwork node, a second network tending to forward-bias saidprotective transistor base-emitter junction in response to the magnitudeof the load current flowing through said resistive means in onedirection with respect to the load, and a third network tending toreverse-bias said protective transistor base-emitter junctionproportional to the magnitude of the voltage swing of said other outputterminal in one direction, said third network being inoperative andproviding no reverse bias when said output terminals areshort-circuited. .Iaddend. .Iadd.
 79. An amplifier as set forth in claim78 wherein said second network includes conductors connecting theprotective transistor base to one end of said current-sensing resistorand the protective transistor emitter to the other end of saidcurrent-sensing resistor, and said third network includes a conductornetwork connecting the protective transistor emitter to said one outputterminal to cause the voltage of said emitter to swing in the directionof the voltage swing of said one output terminal, and further includinga conductive path extending from the protective transistor base to anode at approximately ground potential. .Iaddend. .Iadd.
 80. Anamplifier as recited in claim 76 wherein said D.C. feedback networkcomprises a first resistor, a second resistor and a capacitor, saidfirst resistor having one end direct-current-coupled to said outputstage midpoint and its other end direct-current-coupled to said feedbackinjection node, said second resistor having one end connected to saidother end of said first resistor and its other end connected to one endof said capacitor, the other end of said capacitor being connected toground, whereby said D.C. feedback network injects into said node both aD.C. feedback signal and an A.C. feedback signal with the magnitude ofthe D.C. feedback signal being substantially greater than the magnitudeof the A.C. feedback signal, thereby minimizing the D.C. offset at saidoutput terminals and maximizing the A.C. feedback stability of theamplifier. .Iaddend. .Iadd.
 81. A transistor power amplifier as setforth in claim 80 wherein said output stage comprises a pair of outputtransistors of the same polarity type, each of said output transistorshaving a base, an emitter and a collector, means connecting thecollector of a first of said output transistors to a first of said powersupply terminals, means connecting the emitter of said first outputtransistor to said output terminal, means connecting the collector ofthe second output transistor to said output terminal, means connectingthe emitter of the second output transistor to the second power supplyterminal, means direct-current-coupling the emitter of a first of saiddrive stage transistors to the base of said first output transistor, andmeans direct-current-coupling the collector of the second drive stagetransistor to the base of said second output transistor. .Iaddend..Iadd.
 82. A transistor power amplifier as set forth in claim 81 andcomprising a dissipation-limiting protective circuitincludingcurrent-sensing resistive means in series with said outputterminal and said load, a pair of complementary protective transistorseach having a base-emitter junction and a collector, a pair of diodeseach connecting the collector of a respective one of said protectivetransistors to the amplifier network extending between the collectoroutput of the second common emitter stage and the drive stage, meanstending to forward-bias a respective one of said protective transistorbase-emitter junctions in response to the magnitude of the load currentflowing through said resistive means in a respective direction, a nodemaintained at a potential approximately midway between the potentials ofsaid power supply terminals, and first conductive means connecting theprotective transistor bases to said node and second conductive meansconnecting the protective transistor emitters to said output stagemidpoint so as to apply to a respective one of said protectivetransistor base-emitter junctions a voltage proportional to the voltageswing of the output stage midpoint and tending to reverse-bias saidrespective base-emitter junction. .Iaddend. .Iadd. An amplifier asrecited in claim 76 whereinsaid first common-emitter stage comprisesanother transistor of said predetermined polarity type and having adiode including as base and an emitter, said last-recited emitter beingcoupled to the emitter of said first-recited common-emitter stagetransistor, said feedback signal being transmitted in the feedbackdirection through said diode. .Iaddend. .Iadd.
 84. Adirect-current-coupled amplifier as recited in claim 83 and capable oftransmitting a large low-frequency transient signal such as is generatedwhen a tone arm is dropped upon a record or when switching betweensignal sources or due to other causes, and further comprising adissipation-limiting circuit to prevent the respective operating pointsof the output stage transistors from entering a region of highdissipation in response to said large low-frequency transient signal..Iaddend. .Iadd.
 85. An amplifier as set forth in claim 84 wherein saiddissipation-limiting protective circuit comprises current-sensingresistive means in series with said other output terminal and the saidload and for conducting and sensing the instantaneous load current, apair of protective transistors of complementary polarity and each havinga base-emitter junction and a collector, said second common-emitterstage transistor having a collector output network, first meansconnecting one of said protective transistor collectors to saidcollector output network of the second common-emitter stage, secondmeans connecting the other of said protective transistor collectors tosaid collector output network of the second common-emitter stage, firstmeans tending to forward-bias one of said protective transistorbase-emitter junctions in response to the magnitude of the load currentflowing through said resistive means in one direction with respect tothe load, second means tending to forward-bias the other of saidprotective transistor base-emitter junctions in response to themagnitude of the load current flowing through said resistive means inthe opposite direction with respect to the load, third means tending toreverse-bias said one protective transistor base-emitter junctionproportional to the magnitude of the voltage swing of said otherterminal in one direction, and fourth means tending to reverse-bias saidother protective transistor base-emitter junction proportional to themagnitude of the voltage swing of said other terminal in the oppositedirection. .Iaddend. .Iadd.
 86. An amplifier as set forth in claim 85and comprisinga node maintained at approximately ground potential, saidthird and fourth reverse-bias means including resistive means connectingsaid protective transistor bases to said node. .Iaddend. .Iadd.
 87. Anamplifier as set forth in claim 86 wherein each of said first and secondconnecting means comprises a diode connected between the respectiveprotective transistor collector and said second common-emitter stagecollector output network. .Iaddend. .Iadd.
 88. An amplifier as recitedin claim 76, said first common-emitter transistor having a baseelectrode and an emitter electrode and having a quiescent base-emitterpotential difference between said electrodes, said base bias networkresistor constituting a first resistive means associated with said baseelectrode and having a first quiescent voltage thereacross, saidcenter-tap being substantially at ground potential, a second resistivemeans associated with said emitter electrode and constituting at leastpart of said feedback network and having one end connected to said otheroutput terminal and having a second quiescent voltage thereacross,network means for maintaining the quiescent potential of one of saidelectrodes at a level displaced from ground potential by an amountsubstantially equal to the sum of said quiescent base-emitter potentialdifference and said quiescent voltage across the resistive meansassociated with said other network electrode to maintain the quiescentvoltage of the output at substantially ground potential, and means fordirect-current coupling the other end of said second resistive means tosaid emitter electrode to transmit said feedback signal to the latter inthe back direction and the voltage across said coupling means being of amagnitude to enable said other output terminal quiescent voltage to bemaintained at substantially the same ground potential as said one outputterminal. .Iaddend. .Iadd.
 89. An amplifier as recited in claim 88wherein said last-recited direct-current-coupling means comprises adiode junction, said first common-emitter stage comprising a secondtransistor having a base-emitter junction constituting said diodejunction and having a base and an emitter coupled to the emitter of thefirst-recited transistor of the first common-emitter stage, said secondresistive means being connected to said base of said second transistorof the first common-emitter stage. .Iaddend..Iadd.
 90. An amplifier asrecited in claim 88 wherein said last-recited direct-current-couplingmeans comprises a direct conductive connection of the other end of saidsecond resistive means to said emitter electrode. .Iaddend. .Iadd.
 91. Atransistor power amplifier for high-fidelity music reproduction withoutaudible distortion or listening fatigue under extended criticallistening conditions and comprising a first amplifier stage including aninput circuit and an output circuit, an output stage having a firstoutput terminal for direct-coupling to one terminal of a loudspeaker orother load, a power supply connected to said output stage, a secondoutput terminal for direct-current-coupling to the other terminal ofsaid load, first circuit means direct-current-coupling said firstamplifier stage output circuit to said output stage to drive the latter,a substantially fixed source of reference potential of a magnitudesubstantially independent of variations in the potential of said powersupply, second circuit means connected to said reference potentialsource and said first amplifier stage input circuit to supply biascurrent thereto at a relatively fixed quiescent potential so as toadjust the quiescent potential of said first output terminal to that ofsaid second output terminal, a node having a direct-current potentialvarying substantially proportionately to that of said first outputterminal, and a passive feedback impedance direct-current-coupledbetween said node and said first amplifier stage, thereby providing adirect-current negative feedback loop to reduce any tendency of thequiescent potential of said first output terminal to vary from that ofsaid second output terminal, said first amplifier stage comprising adifferential pair of transistors including a first transistor and asecond transistor, said first transistor being connected in thecommon-emitter node and having a collector forming part of said firstamplifier stage output circuit whereby the signal at said collector istransmitted to said output stage, said first transistor having a baseforming part of said first amplifier stage input circuit, a groundconstituting said reference potential source, said second circuit meansincluding a resistor connected between said ground and said firsttransistor base, said first and second transistors having mutuallyconnected emitters, said second transistor having a base, said passivefeedback impedance being direct-current-coupled to said secondtransistor base, said output stage including at least a pair of outputtransistors connected in a push-pull arrangement and each having a base,said first circuit means including a push-pull drive stage having atleast two complementary transistors each direct-current-coupled to arespective one of said output transistor bases, said drive and outputstages being connected to operate in the emitter-follower mode..Iaddend. .Iadd.
 92. An amplifier as recited in claim 91 whereinsaidsecond output terminal is directly connected to said ground, a secondcommon-emitter transistor having a base direct-current-coupled to saidfirst transistor collector and having a collector, said secondcommon-emitter transistor being of a polarity type opposite to that ofsaid first transistor, each of said two drive stage transistors having abase, third circuit means direct-current-coupling said secondcommon-emitter transistor collector to said drive stage transistorbases, and means for transmitting an input signal to said firsttransistor base. .Iaddend. .Iadd.
 93. A transistor power amplifier forhigh-fidelity music reproduction without audible distortion or listeningfatigue under vigorously critical extended listening conditions andcomprising a single-ended push-pull output stage having a first outputterminal for direct-current-coupling said stage to one end of aloudspeaker or other load, a power supply connected to said outputstage, a second output terminal for direct-current-coupling the otherend of said load to said power supply, a preceding amplifying stagecomprising a transistor having a base electrode and an emitterelectrode, an active network direct-current-coupling said precedingstage to said output stage, a bias network maintaining one of saidelectrodes at a relatively fixed direct-current quiescent potentialindependent of variations in the load current or in the power supplypotential, and a negative feedback network responsive to the potentialof said first output terminal and direct-current-coupled to the otherelectrode for applying thereto a potential proportional to thedirect-current quiescent potential of said first output terminal tomaintain said quiescent potential substantially equal to that of saidsecond output terminal, and wherein said preceding amplifying stagetransistor is of a predetermined polarity type and connected in thecommon-emitter mode so as to constitute a first common-emittertransistor, said active network including a second common-emitter stagehaving only a single amplifying transistor of a polarity type oppositeto that of said first common-emitter transistor and having an emitterand a collector output network, a push-pull drive stage including a pairof complementary transistors each having a base, an emitter and acollector, said second common-emitter transistor collector outputnetwork being direct-current-coupled to said bases of the drive stagetransistors, said power supply having positive and negative terminals,conductive means connecting said emitter of said second common-emitterstage transistor to one of said supply terminals to maintain thepotential of said emitter at approximately the potential of said supplyterminal, means connecting said drive stage transistor collectors tosaid power supply terminals respectively, means direct-current-couplingsaid drive stage to said output stage, means connecting said drive stagetransistor emitters to the output terminal whereby said drive and outputstages operate in the emitter-follower mode, said negative feedbacknetwork transmitting a feedback signal to said emitter electrode of saidpreceding amplifying stage transistor, said bias network including aresistor having one end connected to said base electrode and its otherend connected to ground, said feedback network including a feedbackresistor for transmitting a feedback signal in the back direction towardsaid last-recited emitter electrode, and conductive means separate fromsaid feedback resistor and connected to said emitter electrode forconducting at least a substantial portion of the emitter current of saidfirst common-emitter preceding amplifying stage transistor. .Iaddend..Iadd.
 94. A transistor power amplifier as set forth in claim 93whereinsaid output stage comprises a pair of output transistors of thesame polarity type, each of said output transistors having a base, anemitter and a collector, means connecting the collector of a first ofsaid output transistors to a first of said power supply terminals, meansconnecting the emitter of said first output transistor to said outputterminal, means connecting the collector of the second output transistorto said output terminal, means connecting the emitter of the secondoutput transistor to the second power supply terminal, meansdirect-current-coupling the emitter of a first of said drive stagetransistors to the base of said first output transistor, and meansdirect-current-coupling the collector of the second drive stagetransistor to the base of said second output transistor. .Iaddend..Iadd.
 95. An amplifier as set forth in claim 93 wherein said outputstage comprises a pair of output transistors of the same polarity type,each of said output transistors having a base, an emitter and acollector, means connecting the collector of a first of said outputtransistors to a first of said power supply terminals, means connectingthe emitter of said first output transistor to said output terminal,means connecting the collector of the second output transistor to saidoutput terminal, means connecting the emitter of the second outputtransistor to the second power supply terminal, a push-pullemitter-follower second drive stage comprising a pair of transistors ofthe same polarity type as said output transistors and each having acollector, a base and an emitter, means direct-current-coupling theemitter of a one of said first-recited drive stage transistors to thebase of one of said emitter-follower drive stage transistors, meansdirect-current-coupling the collector of the other of said first-reciteddrive stage transistors to the base of the other of saidemitter-follower drive stage transistors, and meansdirect-current-coupling the emitter of each of said emitter-followerdrive stage transistors to the base of a respective one of said outputtransistors. .Iaddend. .Iadd.
 96. An amplifier as recited in claim 93whereinsaid first common-emitter stage has a high-frequency roll-off ata predetermined frequency, said second common-emitter stage having ahigh-frequency roll-off at a second frequency substantially lower thansaid predetermined frequency, whereby a large amount of feedback may beutilized with stability because of the low-frequency stability marginprovided by the direct-coupling, in cooperation with the high-frequencystability margin provided by the relative respective high-frequencyroll-offs of the first and second common-emitter stages. .Iaddend..Iadd.
 97. An amplifier as recited in claim 96 wherein saidhigh-frequency roll-off of said second common-emitter stage is providedby a capacitor connected between the collector and base of the secondcommon-emitter stage transistor. .Iaddend..Iadd.
 98. An amplifier asrecited in claim 96 wherein said drive stage comprises a pair ofcomplementary transistors each having an emitter connected to saidoutput terminal whereby the drive and output stages operate in theemitter-follower mode so as to increase said high-frequency stabilitymargin. .Iaddend. .Iadd.
 99. A pair of amplifiers each as recited inclaim 93, each of said amplifiers having individual input and outputterminals so as to constitute an independent channel, means forconnecting said two amplifiers either in a separate mode or in a seriesmode, means for connecting the output terminals of one amplifier to afirst loudspeaker and the output terminals of the other amplifier to asecond loudspeaker when said amplifiers are connected in said separatemode, means for transmitting a first input signal to the input of saidone amplifier and a second input signal to the input terminals of saidother amplifier when said amplifiers are connected in said separatemode, means for connecting an output terminal of one amplifier to oneterminal of a loudspeaker and an output terminal of the other amplifierto the other terminal of the same loudspeaker when said amplifiers areconnected in said series mode, and means for transmitting an inputsignal to the input terminals of said one amplifier and anoppositely-phased replica of said input signal to the input terminals ofsaid other amplifier when said amplifiers are connected in said seriesmode. .Iaddend. .Iadd.
 100. A transistor power amplifier forhigh-fidelity music reproduction without audible distortion or listeningfatigue under extended critical listening conditions and comprising asingle-ended push-pull output stage having a first output terminal fordirect-current-coupling said stage to one end of a loudspeaker or otherload, a power supply connected to said output stage, a second outputterminal for direct-current coupling the other end of said load to saidpower supply, a preceding amplifying stage comprising a transistorhaving a base electrode and an emitter electrode, an active networkdirect-current-coupling said preceding stage to said output stage, abias network maintaining one of said electrodes at a relatively fixeddirect-current quiescent potential independent of variations in the loadcurrent or in the power supply potential, and a negative feedbacknetwork responsive to the potential of said first output terminal anddirect-current-coupled to the other electrode for applying thereto apotential proportional to the direct-current quiescent potential of saidfirst output terminal to maintain said quiescent potential substantiallyequal to that of said second output terminal, and whereinsaid precedingamplifying stage transistor is connected in the common-emitter mode,said active network including a push-pull drive stage comprising a pairof complementary drive transistors each having an emitter and a base,means connecting said emitters to said output terminal whereby saiddrive and output stages together effectively operate in theemitter-follower mode, said preceding common-emitter amplifying stagetransistor having a collector and an emitter and being of apredetermined polarity type, a second common-emitter amplifying stageincluding a transistor of a polarity type opposite to that of saidpreceding amplifying stage transistor and having a base and a collector,said preceding common-emitter transistor collector beingdirect-current-coupled to said second common-emitter transistor base,network means direct-current-coupling said second common-emittertransistor collector to said drive stage transistor bases, said biasnetwork maintaining said base electrode of said first common-emitterpreceding amplifying stage transistor at said relatively fixeddirect-current quiescent potential and including a resistor extendingfrom said base electrode to ground, said negative feedback networktransmitting a direct-current feedback signal to said emitter electrodeof the first common-emitter preceding amplifying stage transistor, saidfeedback network comprising a first resistor, a second resistor and acapacitor, said first resistor having one end direct-current-coupled tosaid output stage midpoint, said second resistor having one endconnected to the other end of said first resistor and its other endconnected to one end of said capacitor, the other end of said capacitorbeing connected to ground, said output stage comprising a pair of outputtransistors of the same polarity type and each having a base and anemitter, one of said drive transistors having an emitterdirect-current-coupled to the base of a respective one of said outputtransistors, the other of said drive transistors having a collectordirect-current-coupled to the base of the other output transistor,conductive means connecting the emitter of said one output transistor tosaid first output terminal, said power supply having a supply terminalungrounded with respect to D.C., conductive means connecting the emitterof said other output transistor to said ungrounded power supplyterminal. .Iaddend. .Iadd.
 101. A direct-current-coupled amplifier asrecited in claim 100 capable of transmitting a large low-frequencysignal and further comprising a dissipation-limiting circuit to preventthe respective operating points of the output stage transistors fromentering a region of high dissipation in response to said low-frequencysignal,current-sensing resistive means in series with said outputterminals and said load and for conducting and sensing the instantaneousload current, a pair of protective transistors of complementary polarityand each having a base-emitter junction and a collector, meansconnecting each of said protective transistor collectors to a nodepreceding said drive stage, first means tending to forward-bias one ofsaid protective transistor base-emitter junctions in response to themagnitude of the load current flowing through said resistive means inone direction with respect to the load, second means tending toforward-bias the other of said protective transistor base-emitterjunctions in response to the magnitude of the load current flowingthrough said resistive means in the opposite direction with respect tothe load, third means activated when the output terminals are notshorted and tending to reverse-bias said one protective transistorbase-emitter junction proportional to the magnitude of the voltage swingof said first output terminal in one direction, and fourth meansactivated when the output terminals are not shorted and tending toreverse-bias said other protective transistor base-emitter junctionproportional to the magnitude of the voltage swing of said first outputterminal in the opposite direction, a node maintained at approximatelyground potential, said third and fourth reverse-bias means including aconductive path connecting the protective transistor bases to said nodeand means connecting the protective transistor emitters to said firstoutput terminal. .Iaddend. .Iadd.
 102. An amplifier as recited in claim101 wherein said connecting means comprises diodes connecting saidprotective transistor collectors to said network means whichdirect-current-couple said second common-emitter transistor collector tosaid drive transistor bases. .Iaddend.